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BACKGROUND OF THE INVENTION
This invention relates to a static controller for power factor correction
or adaptive filtering, and more particularly to a cascaded polyphase
cycloconverter and single phase high frequency resonant tank circuit
controlled so as to provide distribution system power factor correction,
either leading or lagging, or alternately to function as a fast response
adaptive capacitive filter or inductive filter.
System power factor corrrection in electrical power distribution systems if
often required for various lagging loads, and is normally accomplished by
polyphase capacitor banks operating at distribution frequency. Also,
occasionally a fast system response is required to eliminate damaging
resonances in faulted distribution systems or to sufficiently attenuate
variable amplitude and/or frequency line harmonics caused by switching
type loads. Variable leading reactive power can be obtained by switched
capacitor banks, using phase controlled thyristors connected in series
with inductors to vary the equivalent reactance. The present invention is
directed to an alternate and more versatile solid state circuit that is
lightweight with a fast response and provides either power factor
correction or adaptive filtering. Both applications are quite common, such
as for use with large KVA phase control systems.
The cascaded high frequency link cycloconverter system is disclosed in U.S.
Pat. No. 3,742,336 to B. D. Bedford and in allowed application Ser. No.
419,490 now U.S. Pat. No. 3,882,369 to W. McMurray, both assigned to the
assignee of this invention, and usually is comprised by input and output
cycloconverters separated by a high frequency resonant tank which provides
commutation for both cycloconverters. As typically used with an inductive
load, the tank frequency is variable to control the amount of reactive
power, since the capacitor reactive power increases and the inductor
reactive power decreases as the frequency is raised above the resonant
frequency. When used without the output cycloconverter as here taught, the
high frequency link converter distinguishes in one manner in that only the
reactive power needed for power factor correction of filtering is required
and not the full load power.
SUMMARY OF THE INVENTION
In accordance with the invention, a static reactive power controller based
on the high frequency link cycloconverter approach utilizes essentially
only a polyphase cycloconverter in cascade with a single phase high
frequency resonant tank circuit usually comprised by a single inductor and
capacitor in parallel, the cycloconverter further having an input filter
including in each phase a series filter inductor which is connected to a
source of low frequency line or distribution voltage. Control means are
provided for controlling the cycloconverter to produce real power flow to
the resonant tank only approximately sufficient to supply the losses,
while producing a variable amount of leading or lagging reactive power as
determined by the polarity and magnitude of a control signal
representative of a system quantity to be controlled. Depending on the
application, the control signal is a distribution system power factor
correction signal, a VAR control signal, etc. Preferably the real power is
controlled by a tank voltage error signal which functions to maintain an
approximately constant tank voltage. Thus, the phase and amplitude shifter
in the control circuit generates cycloconverter reference signals for
effectively controlling the real and reactive power of the cycloconverter
as determined respectively by the tank voltage error signal and the
externally generated control signal.
Since leading as well as lagging reactive power can be controlled in
vernier fashion with a fast response, the static controller can provide an
electronically variable capacitance in the previously mentioned
applications for power factor correction, adaptive filtering, and VAR
control. Thus, the usual polyphase capacitor acting at distribution
frequency is replaced by a single phase high frequency capacitor thereby
reducing filter size and improving system response.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a simplified block diagram of the cycloconverter system for
variable power factor correction or filtering with a smaller high
frequency capacitor;
FIG. 2 is a schematic circuit diagram partially in block diagram form of
the cycloconverter system and one embodiment of a suitable control
circuit;
FIG. 3 is a system block diagram showing only a single phase of the control
circuit in greater detail than in FIG. 2;
FIG. 4 shows diagrammatically a cycloconverter analogy to a synchronous
machine;
FIGS. 5-8 are phasor diagrams illustrating the theoretical basis for real
power control and input power factor control;
FIG. 9 is a block diagram of a phase and amplitude shifter for generating
cycloconverter reference signals for independently controlling real power
and input power factor using either polarity of d-c control signals;
FIG. 10 is a phasor diagram useful in explaining operation of the static
controller for power factor correction and adaptive filtering; and
FIG. 11 is a phasor diagram useful in explaining operation of the different
form of phase and amplitude shifter shown in FIGS. 2 and 3.
DESCRIPTION OF THE PREFERRED EMBODIMENT
The high frequency link static reactive power controller as shown in FIG. 1
is most commonly connected to a polyphase voltage distribution system and
can be controlled to produce either leading or lagging reactive power in
typical applications such as power factor correction, VAR control, and
adaptive filtering. The static controller is comprised by a polyphase
ac/ac cycloconverter 15 connected in cascade with a high frequency
resonant tank circuit 16 or parallel resonant L-C commutation circuit
preferably including only a single inductor 17 and capacitor 18 in
parallel. The amount and sign of the reactive power produced by the static
controller is determined by an externally generated control signal
v.sub.s1 of either polarity that is representative of a system quantity to
be controlled. The control signal v.sub.s1 for example is a system power
factor correction signal or a VAR control signal generated by a device
which includes a power factor or VAR sensor 19 for sensing the appropriate
distribution system parameters and deriving a signal proportional to the
difference from a desired value. The preferred embodiment is explained
with regard to power factor correction of a system with a lagging power
factor load 20 energized by a three-phase, 60 Hz or other low frequency
voltage source. Of course, usually a unity system power factor is desired
although the static controller herein described has a fast response and
regulates power factor as desired. As compared to the prior art polyphase
capacitor operating at distribution frequency, power factor correction in
a polyphase system is achieved with a single phase, electronically
variable, high frequency capacitor, thereby reducing filter size and
improving system response.
The static reactive power controller can be constructed in either
delta-connected or wye-connected versions similar to the high frequency
link cycloconverter systems, assuming that the output cycloconverter and
output filter are not used, illustrated in FIGS. 14 and 15 of the
previously mentioned U.S. Pat. No. 3,882,369, to which the reader may
refer for further information. The preferred wye-connected cyclo-converter
system here shown in FIG. 2 uses a twelve-thyristor cycloconverter 15
comprised by three single phase-to-single phase cycloconverters. In each
single phase cycloconverter, one pair of inverse-parallel thyristors 21 is
connected to one junction of the tuned high frequency resonant tank 16
while the other pair of thyristors is connected to the other junction, and
both pairs are connected through a series reactor or filter inductor 22 to
one of the input terminals 23. The input filter also includes three filter
capacitors 24 provided between the respective input terminals 23 and the
neutral terminal N, the neutral terminal also being connected to the
center tap of tank inductor 17. As most commonly used, the source is a
three-phase, 60 Hz, 230 volts supply, and the high frequency tank is
operated at about 2-4 kHz. Cycloconverter 15 operates in the regenerative
or inverting mode and drives the tuned high frequency tank 16 directly
from the polyphase ac supply at a frequency above its resonant frequency,
the amount of commutating energy for the thyristors being determined by
the tank frequency. The cycloconverter control circuit or control means
permits either direction of power flow through the cycloconverter,
however, and in the event of power flow from the tank to the source the
cycloconverter is controlled in conventional fashion to operate in
rectifying mode.
In accordance with the invention, the control circuit shown at the bottom
of FIG. 2 controls the cycloconverter to produce real power flow to the
resonant tank only approximately sufficient to supply the static
controller losses, i.e., there is essentially no real power flow to the
resonant tank. Also, as was previously mentioned, the cycloconverter is
controlled to selectively produce a variable amount of leading and lagging
reactive power as determined by the polarity and magnitude of the control
signal v.sub.s1. A tank voltage error signal is used as the real power
control signal v.sub.s2 and maintains the high frequency tank voltage
constant or within predetermined limits, in view of the fact that the
resonant tank 16 is a relatively small reservoir of energy as compared to
the system as a whole. Too high a tank voltage can damage the components,
while too low a tank voltage can result in excessive input currents for
much the same reason as when the back emf of a synchronous machine is low.
In the synchronous machine analogy, to be explained in detail later, and
using phase A by way of illustration, the phasor summation of the per
phase sine wave line voltage E.sub.b at the input terminal 23 and the
voltage produced across the filter inductor 22 is the induced voltage
E.sub.s of the cycloconverter. In the voltage control system to be
described, the cycloconverter control circuit establishes the phase and
amplitude of the induced voltage E.sub.s so that the static controller
produces the commanded amount of reactive power while maintaining the high
frequency tank voltage within prescribed limits for either direction of
power flow. With respect to the distribution system line voltage used as a
reference, the sign and amplitude of the reactive power (or the input
power factor) are respectively determined by the sign and amplitude of the
direct component of voltage, while the direction and amount of real power
flow are respectively determined by the sign and amplitude of the
quadrature component of voltage. The control circuit to be described
actually controls the input power factor, however the predominant effect
since the real power flow is small is to control the magnitude of the
reactive power. The basis of operation of the static controller using the
voltage control system is that the cycloconverter is controlled to produce
essentially no real power flow to the resonant tank while generating an
induced voltage E.sub.s approximately in phase with the distribution
system line voltage, with the result that the line current is
approximately 90.degree. displaced from a distribution system line
voltage. As is evident in FIG. 3 to those skilled in the art, the per
phase cycloconverter reference signals E.sub.s-ref to satisfy this control
scheme and mode of operation are employed in conventional fashion in the
cycloconverter firing and control circuit, using the cosine firing wave
phase control technique, to generate appropriately timed firing pulses for
the thyristor switches.
Before proceeding further, the synchronous machine analogy and the
theoretical basis for the construction and operation of the phase and
amplitude shifter, an essential component of the cycloconverter control
means, will be explained with regard to FIGS. 4-9. In the diagram shown in
FIG. 4 of a cycloconverter analogy to a synchronous machine operating on
infinite bus or from a "stiff" source, 60 Hz power is supplied through
reactor X.sub.s (i.e., filter inductor 22) to the cycloconverter. For
simplicity the impedance X.sub.s is assumed to be purely reactive, and
E.sub.b and E.sub.s are as previously identified in FIG. 2. Following the
synchronous machine analogy, I.sub.c is the current flowing in a direction
from the cycloconverter to the source, and I.sub.L = -I.sub.c is the
current flowing from the source to the cycloconverter.
Referring now to FIG. 5 which gives the phasor diagram for real power
control, the induced voltage phasor E.sub.s can be constructed as
E.sub.s = E.sub.b + E.sub.z = E.sub.b + E.sub.x + E.sub.R,
where
E.sub.z = IX.sub.s, E.sub.x = I.sub.x X.sub.s and E.sub.R = I.sub.R
X.sub.s.
Therefore, when the E.sub.z phasor is constructed at a fixed angle .phi.
and its amplitude is modulated as illustrated in FIG. 5, the real power
will be modulated maintaining the input power factor constant. It will be
noted that FIG. 5 is drawn with respect to the line current I.sub.L, in
which case the angle .delta. between E.sub.b and E.sub.s is less than zero
and power flow is from the source to the cycloconverter. As shown in FIG.
6, when E.sub.z is constructed in the first quadrant by reversing E.sub.R,
the cycloconverter will revert from the inverting or regenerative mode to
the rectifying or active mode, and real power flows in the other direction
from the cycloconverter to the source. It is noted in FIG. 6 that the
angle .delta. is greater than zero and that the powr flow reverses as
compared to FIG. 5 since I.sub.L is constructed in the second rather than
in the first quadrant.
FIG. 7 shows the phasor diagram for leading input power factor control for
source to cycloconverter power flow. To modulate the input power factor,
conventionally defined as cos .phi., the phasor E.sub.x is modulated to
control the input power factor while maintaining the real power constant.
FIG. 8 applies to lagging power factor control for source to
cycloconverter power flow. The phasor E.sub.x changes sign as compared to
FIG. 7 and is modulated, and therefore the power factor changes from the
leading to the lagging region. In this regard it is observed that the line
current I.sub.L changes from the first to the fourth quadrant. In all of
these diagrams the angle .epsilon. between E.sub.b and E.sub.s is assumed
to be small.
FIG. 9 shows a simple phase and amplitude shifter 26 for generating single
phase cycloconverter reference voltage signals which can control the real
power and input power factor of the cycloconverter independently and
linearly by dc signal voltages. The phase shifter circuit is insensitive
to bus or line voltage frequency drift, is distortion free, and has an
almost instantaneous response characteristic. In addition, the real power
and power factor angle can be changed to either polarity by simply
reversing the polarity of the dc control signals. Of course, for a three
phase shifter three of the single phase circuits shown in FIG. 9 are
required. Ordinarily, the signal levels are reduced by the gain factor of
the cycloconverter, and thus the per phase input sine wave line voltage
signal E.sub.b is preferably obtained in the case of the FIG. 2 system by
means of a step-down potential transformer directly coupled between one
input terminal 23 and the neutral terminal N. The input line voltage
signal (see FIG. 9) is converted to a cosine wave of proportional
amplitude using a frequency insensitive sine-cosine converter 27, such as
the device described in the copending application Ser. No. 561,592 by B.
K. Bose and the inventor, entitled "Frequency Insensitive Sine
Wave-to-Cosine Wave Converter," filed on Mar. 24, 1975, and assigned to
the same assignee as this invention. This converter employs the
trigonometric relationship cos .omega.t = .sqroot.1-sin.sup.2 .omega.t and
is operative over a wide frequency range with an almost instantaneous
response characteristic to produce cosine waves with an amplitude
proportional to the sine wave amplitude. In one form implemented with
presently available integrated circuits, the converter includes an analog
multiplier for generating a sine wave squared signal, a clamping circuit
for effectively shifting the voltage level, a sign inverter for generating
a cosine wave squared signal, a square rooter for producing a
negative-going full wave rectified cosine wave, a second sign inverter,
and a polarizer for converting the positive-going full wave rectified
cosine wave to the desired ac cosine wave. Another suitable prior art
technique for frequency insensitive cosine wave generation involves
integrating the sine wave and then multiplying the amplitude of the cosine
wave by a voltage proportional to the frequency. The dc real power control
signal v.sub.s2 is used as a polarity reversing signal for sine-cosine
converter 27, and thus when v.sub. s2 is negative a negative cosine wave
is generated. In a parallel branch, E.sub.b is fed to an analog
four-quadrant multiplier 28 used as a variable gain amplifier, the second
input to the multiplier being the dc power factor control signal v.sub.s1
which can be of either polarity depending upon whether a leading or
lagging power factor is desired. The resulting variable amplitude sine
wave has a peak amplitude and polarity depending upon the magnitude and
polarity of v.sub.s1, and is combined with the frequency insensitive
cosine wave in summer 29 to generate at its output a summation signal. In
terms of the phasor diagram for power factor control in FIG. 7, the
in-phase sine wave signal (indicative of E.sub.x) is multiplied or
modulated according to the desired power factor, while the cosine wave
signal (indicative of E.sub.R) remains constant, the two being summed
preferably using an operational amplifier to obtain the summation signal
(indicative of E.sub.z).
To implement the real power control, the output of summer 29 is fed to an
analog two-quadrant multiplier 30, the second input to this multiplier
being the absolute value of the real power control signal v.sub.s2. A
circuit 31 is used to invert the negative polarity dc signal since
v.sub.s2 is of either polarity depending upon the commanded direction of
power flow through the cycloconverter. In terms of the phasor diagram for
real power control given in FIG. 5, the effect of using multiplier 30 is
to vary the amplitude of the input summation signal according to the
magnitude of v.sub.s2. As the final step, the signal -E.sub.b produced by
sign inverter 32 is combined with the variable amplitude summation signal
representative of -E.sub.z using a second operational amplifier summer 33.
The output is the desired phase and amplitude shifted cycloconverter
reference voltage E.sub.s-ref. It is obvious that the input power factor
and real power can be electronically adjusted independently of one
another, and that either may be held constant while varying the other.
With the foregoing discussion of FIGS. 4-9 as background, the theoretical
basis for operating the static controller to generate leading reactive
power for power factor correction can be explained. In FIG. 10, the phasor
E.sub.R is constructed in the fourth quadrant and the angle .delta. is
relatively small and less than zero for source to cycloconverter power
flow. For the three cases illustrated, E.sub.R remains constant as E.sub.x
is modulated with a consequent change in the induced voltage E.sub.s.
Although there is a change in the power factor angle .phi., the change is
relatively small and the predominant change is in E.sub.x. The line
current I.sub.L is constructed in the first quadrant and leads the
distribution system line voltage E.sub.b by nearly 90.degree.. In similar
fashion, I.sub.R and .phi. show no or little change and the predominant
change is in I.sub.x, which leads E.sub.b by exactly 90.degree.. When
E.sub.x is reversed in direction and modulated, the phasor I.sub.x is
reversed in direction and constructed in the fourth quadrant and the line
current I.sub.L then lags the line voltage E.sub.b by 90.degree.. These
form the basis for leading and lagging reactive power control in the
voltage control system, assuming that only sufficient real power is
supplied to the resonant tank to compensate for the inherent circuit
losses.
FIG. 2 shows the complete phase and amplitude shifter circuit 26' for a
balanced system with provision for real power control and input power
factor (reactive power) control in either the leading or lagging
direction. Phase shifter 26' is a modification of phase shifter 26 shown
in FIG. 9, which can also be used in the practice of the invention when
there is an unbalanced system. As a special case when the three phase
power supply is balanced in amplitude and phase, the cosine wave or
frequency insensitive E.sub.b < 90 phasor can be generated conveniently by
the addition and subtraction of the phase voltages. FIG. 11 gives the
phasor diagram used to explain the basis of operation of the phase shifter
circuit in FIG. 2 (also see FIG. 3). To obtain the frequency insensitive
cosine wave for each phase, as well as each single phase line voltage
three step-down transformers 35a, 35b and 35c have their primary windings
respectively connected between each input line and neutral. Each single
phase transformer has a split secondary winding such that one gives the
positive polarity phase voltage while the other gives the negative
polarity phase voltage, e.g., E.sub.C and -E.sub.C. The circuit for
producing the phase A cycloconverter reference signal E.sub.sA will be
explained by way of illustration, the other two phases being similar so
that corresponding components in the three phases are indicated by
corresponding numerals. To obtain the frequency insensitive 90.degree.
phase shifted cosine wave (in FIG. 11, see the phasor -E.sub.1 which is
perpendicular to the reference line voltage phasor E.sub.A), the
appropriate secondary windings of transformers 35b and 35c are connected
to obtain a voltage signal representing E.sub.C -E.sub.B which by phasor
addition gives -E.sub.1. To this end, as is illustrated, the dot end of
the appropriate secondary winding of transformer 35b is grounded while the
other end is connected to the undotted end of the appropriate secondary
winding in transformer 35c. The dot or positive polarity end of this
latter secondary winding is coupled directly to a potentiometer 36c for
deriving at the wiper a proportional voltage to be supplied as one input
to the analog four-quadrant multiplier 37a. The positive or negative
polarity real power control signal v.sub.s2 is the high frequency tank
voltage error signal representing the difference between actual and
desired values of tank voltage, and is the second input to multiplier 37a
so as to generate at its output a cosine wave with a modulated amplitude
representing -E.sub.A '. This cosine wave with a polarity and peak
amplitude dependent upon the polarity and magnitude of the real power
control signal is fed through an input resistor to the summing junction
38a of an operational amplifier 39a connected as a summing amplifier. To
obtain input power factor (reactive power) control, the negative-going
sine wave line voltage -E.sub.A is an input to a second two-quadrant
multiplier 40a used as a variable gain amplifier, the other input to
multiplier 40a being the power factor correction or VAR control signal
v.sub.s1. For a gain = 1, PF = 1; while for a gain greater than 1 the PF
is leading; and for a gain between 0 and 1 the PF is lagging. Since the
angle .delta. is small, using multiplier 40a to modulate the magnitude of
the -E.sub.A phasor effectively produces a like change in -E.sub.sA and
thus also in the quadrature component of line current I.sub.x. The
variable amplitude sine wave at the output of multiplier 40a is the other
input to summing junction 38a. Due to the inverting characteristic of
summing amplifier 39a, the cycloconverter reference signal generated at
its output is the desired positive polarity signal E.sub.sA. The
respectively 120.degree. displaced cycloconverter reference signals
E.sub.sB and E.sub.sC are obtained at the outputs of the other channels.
For further information on the amplitude and phase shifter, the reader may
refer to the copending Bose and Espelage application Ser. No. 562,338
filed on Mar. 25, 1975, entitled "Phase Shifter for Controlling the Power
Components and Power Factor of a Cycloconverter," assigned to the same
assignee. In this application FIG. 8 shows a device for independent real
and reactive power control which may have utility in this invention.
A tank voltage feedback circuit is used for generating the tank voltage
error signal indicative of the difference between a sensed instantaneous
tank voltage and a reference tank voltage. A suitable voltage sensor is
used to sense the high frequency tank voltage, such as the potential
transformer 42 shown in FIG. 2 coupled across the resonant tank components
17 and 18 and having a grounded center tap secondary winding the opposite
ends of which are connected to alternately conductive diodes 43 to produce
a full wave rectified ac voltage. A corresponding dc voltage or sensor
signal is generated as by using a resistor-capacitor peak detector 44, and
is fed through an input resistor to the summing junction 45 of an
operational amplifier 46 connected as a summing amplifier. The other input
to the summing junction 45 is a preselected dc reference tank voltage
V.sub.T-ref. A tank voltage error signal generated at the output of
operational amplifier 46 is used directly as the real power control signal
v.sub.s2 and is either positive or negative depending upon the magnitude
of the sensed actual tank voltage as compared to the reference tank
voltage. Thus, the direction of power flow through the cycloconverter 15
changes depending upon the polarity of the tank voltage error signal,
although usually power flow is in the direction from the source to the
cycloconverter. As was mentioned, the cycloconverter firing and control
circuitry 47 actuated by the three per phase cycloconverter reference
signals is of conventional design and is illustrated in somewhat more
detail in FIG. 3.
In FIG. 3, the input filter is shown schematically and partially omitted,
the high frequency resonant tank 16 is shown in its simplest form, and the
static controller losses are illustrated as an equivalent load. Only a
single phase of the phase and amplitude shifter 26' is illustrated
schematically, as is the three-phase transformer 35' for generating the
input sine wave line voltage signals and the 90.degree. phase shifted
cosine wave signals. The main components of the tank voltage control loop
for generating the real power control signal are designated by the same
numerals. To further explain the cycloconverter firing and control
circuitry, the appropriately phase and amplitude shifted cycloconverter
control signal E.sub.s-ref, which commands the induced voltage E.sub.s to
be produced, is compared with a per phase sensed induced voltage E.sub.s
using a summer 49 to derive an error voltage for driving the
cycloconverter firing circuits. In order to use the cosine firing wave
phase control method of control, the instantaneous tank voltage signal
V.sub.T derived from the potential transformer 42 or a similar transformer
is fed to a frequency insensitive cosine firing wave generator 50
preferably of the type described in the previously mentioned application,
Ser. No. 561,592. In the modulator 51, as is known in the art, the
successive intersections of the induced voltage error signal with the high
frequency cosine firing waves are continuously determined to time the
generation of SCR firing pulses by the firing and lock-out circuitry 52.
Depending upon the commanded direction of power flow through the
cycloconverter, one SCR in each inverse-parallel pair is locked out
according to the sign of the induced voltage error signal. For further
information on thyristor firing control using the cosine firing wave
technique, the reader is referred to the book, "The Theory and Design of
Cycloconverters" by William McMurray, The MIT Press, Cambridge, Mass.,
copyright 1972, Library of Congress catalog card No. 70-178121.
With reference to the voltage control system described herein for
controlling the cycloconverter, reference may be made to the concurrently
filed allowed Bose and Espelage application, Ser. No. 573,373 entitled
"Voltage Control System for High Frequency Link Cycloconverter," assigned
to the same assignee. By way of summary, this technique controls the
induced cycloconverter voltage E.sub.s to be approximately in phase with
the distribution system line voltage E.sub.b, as a result of which the
line current is approximately 90.degree. leading or lagging with respect
to the line voltage. The magnitude and sign of the line current depends
upon the magnitude and sign of the voltage difference between E.sub.b and | | |
