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The invention relates to an electric circuit arrangement for receiving one
of the sidebands of a double sideband signal, comprising two first mixing
stages in which the double sideband signal is mixed with either one of two
locally produced radio frequency oscillations which are shifted in phase
over 90.degree. relative to one another but are of the same frequency, the
output signals of the two mixing stages being filtered by means of first
and second low-pass filters, amplified and supplied to two second mixing
stages in which they are each mixed with either one of two locally
produced low-frequency oscillations shifted in phase over 90.degree.
relative to one another but which are of the same frequency, the output
signals of the two second mixing stages being applied to an adder
(sub-tractor) circuit.
Such a circuit arrangement is known--especially in connection with a
receiver for single sideband signals (cf., for example, DT-AS No. 16 16
312 and "Proc. IRE 44" (1956)pages 1703 to 1705). The frequency of the
locally produced radio frequency oscillation corresponds to the medium
frequency of the received sideband and the frequency of the locally
produced low-frequency oscillation corresponds to the medium frequency of
the useful signal spectrum (third method). With this circuit there is the
risk that the high frequency oscillation is radiated and will cause
squeaking in adjacent receivers. The second mixing stages in which the
low-pass-filtered output signal of the preceding mixing stages is mixed
with the low frequency oscillation and which are, for example, constituted
as balanced modulators, must be of a very symmetrical construction as
otherwise a distortion is produced in the centre of the useful frequency
band owing to mixing with the low frequency oscillation. External
screening and symmetry measures are therefore required, which partially
offset again the advantages of this circuit arrangement (i.a. the fact
they can be easily integrated, as coils are no longer required).
However, the invention relates to a circuit arrangement for receiving
double-sideband signals, particularly with a full or an attenuated
carrier, which can be utilized for example in medium-wave of short-wave
broadcasting.
The prior art circuits generally utilize a frequency conversion for
producing an intermediate frequency and require band filters in the
intermediate frequency range to suppress signals from adjacent channels.
The known receiver circuits can therefore only be partially implemented in
integrated circuit technology and the required filters must be connected
externally. A further drawback of the known circuits consists in
distortions when the so-called "selective fadings" occur. This effect,
which is extremely annoying in radio reception, is produced by the fact
that the carrier changes its amplitude or its phase position relative to
the two sidebands. Consequently, a qualitatively satisfactory reception of
the signals from remote medium-wave or short-wave transmitters is
generally impossible. Also the reception of the image frequency and the
risk of direct reception on the intermediate frequency as well as the
occurrence of spurious beaks and squeaking in the receiving range are
amongst the unpleasant properties of these circuits.
It is an object of the present invention to provide a circuit arrangement
which enables the first-grade reception of double-sideband AM-signals,
also from remote transmitters, and which can be implemented in a simple
manner in integrated circuit technology and which shows no spurious beaks
and squeaking.
Starting from a circuit arrangement of the type mentioned in the preamble
this object is accomplished by the following features:
(a) the frequency of the local radio-frequency oscillation corresponds to
the carrier frequency of the double-side-band signal,
(b) the frequency of the local low-frequency oscillation exceeds or is
equal to the upper frequency of the LF useful signal to be transmitted,
(c) The adder (subtractor) circuit is followed by a third low-pass filter
which suppresses signals having a frequency equal to or above the
frequency of the local low frequency oscillation,
(d) the output signals of the third low-pass filter are mixed in a last
mixing stage with the low-frequency oscillation.
An elaboration of the invention provides that the last mixing stage is
followed by a fourth low-pass filter whose cutoff frequency corresponds to
that of the third low-pass filter. By mixing the output signals of the
third low-pass filter, with the low frequency oscillation in the last
mixing stage, two sidebands are produced, only one of which (the lower
one) can be used. It is true that the frequency of the low-frequency
oscillation can be chosen so that the upper band of these two sidebands is
outside the useful frequency range or outside the audio frequency range
respectively, but the requirements imposed on the third low-pass filter
and which separates the two sidebands of the double-sideband signal would
be much higher.
A further elaboration of the invention ensures that, for the reception of
double-sideband signals with carriers, the radio frequency oscillation is
supplied by a tunable oscillatory circuit, whose frequency is synchronized
by means of a phase and/or frequency control circuit with the
instantaneous carrier frequency. A so-called PLL-circuit (Phase Locked
Loop), which compares the frequency of the oscillator signal with the
frequency of the carrier, comprised in the input signal, and which so
changes the oscillator frequency that the difference of the phase position
between receiver carrier and the locally produced oscillation is brough to
a minimum, can be used as phase or frequency control circuit. This also
requires a low-pass filter having a very small or switcheable cutoff
frequency and a variable control voltage amplifier.
With the circuit arrangement according to the invention only one of the two
sidebands comprised in the double-sideband signal is always used for
reception. So a change in the phase position of the carrier relative to
the two sidebands has no influence on the quality of the reception, which
would manifest itself in a very disturbing manner as selective fading in
the conventional double-side-band receivers. Separating the two sidebands
from one another is effected in the third low-pass filter connected behind
the adder (subtractor) unit. If very low frequencies, for example, 100 Hz
or lower, must be transmitted in the two sidebands, the frequency gap
between these two sidebands is then 200 Hz or less. This frequency gap
also exists at the output of the adder (subtractor) circuit: however, as a
result of the mixing of the double-sideband signal with the radio
frequency and the low frequency oscillation it is moved to the low
frequency range. The third low-pass filter, which separates the two
sidebands, must consequently satisfy very stringent requirements: It must
have a passband corresponding to the bandwidth of the useful signal to be
transmitted (some kHz) and the transition between the passband and the
cutoff range should only be 100 Hz (or less). An elaboration of the
invention therefore provides that this low-pass filter is constituted by a
gyrator filter.
Also the other low-pass filters can be advantageously implemented as
gyrator filters. For the first and second filters which follow the first
radio frequency mixers, there is the additional condition that they must
be fully identical as regards amplitude and also phase response. The phase
of the oscillation presented to said filters by the two mixers is shifted
in one branch 90.degree. relative to the other branch and this phase shift
must be maintained in the filters over the entire frequency range.
Controlled amplifiers must also be built-in in practice which must satisfy,
as far as they are included in the two parallel branches, high
requirements as regards equality of control behaviour. The required
amplitude control voltage (AGC) is obtained by means of a low-pass filter
of a very low cut-off frequency and a self-regulating control voltage
amplifier. Any remaining amplitude errors in the two branches can then be
compensated for by differential gain control in said two branches.
The invention will now be explained in greater detail with reference to an
embodiment shown in the drawing. Herein shows
FIG. 1 a block diagram of a circuit arrangement according to the invention
while FIG. 1a shows a modification of FIG. 1; and
FIGS. 2a to 2g show the frequency spectra in various points of the circuit
arrangement according to the invention.
The double-sideband signal u.sub.O received from the antenna 1 is
applied,--for example via broadband amplifiers, control amplifiers,
attenuators, input circuits etc, which are not shown in the drawing--with
equal phase to two identically constructed parallel branches A and B. FIG.
2a shows the frequency spectrum of the signal u.sub.O. The signal contains
a carrier of the carrier frequency f.sub.T and the two sidebands U and O,
which are arranged symmetrically at the carrier.
Each of the two branches comprises a first mixing stage 1a or 1b, a
low-pass filter 2a or 2b and a further second mixing stage 3a or 3b. In
the mixing stage 1a or 1b the input signal u.sub.O is mixed with a radio
frequency oscillation supplied by a tunable oscillator 4, which
oscillation is applied directly to the mixing stage 1a and via a phase
shifting member 5, which shifts the locally produced oscillation over
exactly 90.degree. to the mixing stage 1b. Preferably the mixing stages 1a
or 1b are formed as full-wave multipliers supplying an output signal which
is proportional to the product of the two input signals. In this ways it
is achieved that the output signals comprise only sideband information
whose frequencies deviate from the receiver carrier frequency.
It is important that the frequency of the oscillation supplied by
oscillator 4 conforms as exactly as possible to the carrier frequency
f.sub.T of the double-side-band signal u.sub.O. To this end a phase or
frequency control circuit--indicated by the block 17, 18--is provided by
means of which the frequency of the locally produced oscillation 4 is
synchronized on the carrier frequency f.sub.T. To control the phase or
frequency, the small positive or negative D.C. voltage which is produced
in branch B in the case of phase errors, can be used. This voltage is
applied via a low-pass filter 17 and the controlled amplifier 18 to the
oscillator 4 and which readjusts the tuning of the oscillator, so that the
phase of the locally produced oscillation corresponds, except for a small
residual error, to the portion of the carrier frequency comprised in the
input signal u.sub.O.
Whilst the positive or negative D.C. voltage occuring in branch B is
proportional to the residual-phase angle .+-. .DELTA. .phi., the D.C.
voltage component in branch A is proportional to the instantaneous
amplitude of the received carrier voltage, and, after having been filtered
by the low-pass filter 19 and gain-controlled in the controlled D.C.
voltage amplifier 20, can be used for amplitude control (AGC).
The D.C. voltage components are then removed by the capacitors 7a and 7b
before the second mixers 3a and 3b.
It is evident that all known measures for linearizing, such as RF-feedback,
high-current transistors, balanced circuits etc. may be used for the
mixers 1a and 1b to avoid cross modulation. Furthermore, by the mixing
process of the input signals with the local high frequency oscillations
the upper and lower sidebands of the received signal are so transposed to
the lower frequency range that the two sidebands are superimposed, one
sideband becoming, as it were, meshed on the other.
However, by mixing the signals with the two oscillator voltages whose
phases are shifted over 90.degree. relative to one another, it is achieved
that the signals in branch B have shifted over +90.degree. for the upper
band signal O and over -90.degree. for the lower band signal U, whereas in
branch A the signals are not shifted and have a 0.degree. phase shift. The
signals u.sub.1 in branch A are therefore shifted over 90.degree. relative
to those in branch B.
In the FIGS. 2b and 2c, which show the signals u.sub.1 and u.sub.1 '
respectively at the output of the low-pass filters 2a and 2b respectively,
this situation is illustrated in that in FIG. 2c the frequency spectrum
originating from the lower sideband u is plotted with a negative sign,
whereas in FIG. 2b the frequency spectrum originating from both sidebands
is indicated with a positive sign.
If, after multiplication in the first mixer, the sideband oscillations O
and U in branch A are cosine functions, then the sideband oscillations
produces in branch B are a sine and a minus-sine function.
This is important for the mixing processes in the second mixers, wherein a
further 90.degree. --shift is effected, so that 180.degree. is obtained.
The low-pass filters 2a and 2b respectively have for their object to
remove the mixing product at or about twice the carrier frequency,
produced at the output of the mixing stages 1a and 1b. This filter
furthermore serves for suppressing the mixing product produced by mixing
the locally produced oscillation with the signal of a transmitter which is
adjacent as regards frequency. If the frequency spacing of the carrier of
two frequency-adjacent transmitters is, for example, 9 kHz--as prescribed
in the plans of the CCIR--the cutoff frequency of the low-pass filter 2a
or 2b must be half this spacing in the assumed example, consequently 4.5
kHz or less, so that the mixing products originating from adjacent
transmitters are suppressed. The low-pass filters 2a or 2b are
consequently used for channel or transmitter separation and they,
consequently, have the same function as the intermediate frequency filter
in a conventional superheterodyne receiver.
At the reception of double-sideband transmissions in which each sideband
contains exactly the same information, the signal u.sub.1 at the output of
the low-pass filter 2a may already represent the low-frequency useful
signal; however, the selective fading already indicated in the preamble,
might be produced when the received carrier is not accurately symmetrical
to the two sidebands or when the locally produced oscillation continuously
changes its phase position relative to the carrier comprised in the input
signal. The effects of the selective fading could only be removed by the
subsequent portion of the circuit.
The output signal of each of the low-pass filters 2a and 2b respectively
are supplied via a controlled amplifier 15 and 16 respectively to either
of a further multiplicative mixing stage 3a and 3b respectively, in which
it is mixed with the low-frequency oscillation produced by an oscillator
8, which oscillation is supplied directly to a mixing stage 3a and to the
other mixing stage 3b through a phase shifting member 9, which shifts the
phase of the locally produced oscillation over exactly 90.degree.. The
low-frequency oscillation supplied by oscillator 8 must have a frequency
f.sub.N, which is equal to or only slightly larger than, the upper
frequency of the useful low-frequency signal to be transmitted; it must
not be chosen too high, as the requirements on the low-pass filter
separating the upper from the lower sideband and which will be explained
in greater detail hereinbelow, are the greater according as the frequency
of the locally produced oscillation is higher. If the frequency spacing
between the carriers of two adjacent transmitters is, for example, 9 kHz,
a value of 4.5 kHz is then preferably chosen in the frequency f.sub.N of
oscillator 8.
The--preferably multiplicative--mixing stages fold the frequency spectra in
accordance with FIG. 2b and 2c respectively symmetrically on frequency
f.sub.N of the added locally produced oscillation so that, for example,
from the frequency band O+U (FIG. 2b), the frequency bands O+U and O'+U'
are produced (cf. FIG. 2d), and from the frequency bands O and U (FIG.
2c), the frequency bands O, U and O', U' (FIG. 2e). Those frequency
components in the two sidebands which correspond to the low useful
frequencies have a smaller frequency spacing from the frequency f.sub.N
than the components which correspond to the higher useful frequencies. The
second mixing operation with the two oscillations, which are shifted
90.degree. relative to one another, of the oscillator 8, furthermore
accomplishes that the upper sideband O appears in the lower frequency band
(frequencies <f.sub.N) in the same phase position at the two outputs of
the mixers 3a and 3b respectively, whereas the lower sideband U appears in
the lower frequency band with the opposite phase at the two outputs of the
mixer stages 3a and 3b (cf. FIG. 2d and 2e).
(The reverse situation is obtained if the phase shifting member 9 is not
arranged between the oscillator 8 and the mixing stage 3b, but between the
oscillator 8 and the mixing stage 3a). In the upper frequency band (i.e.
for frequencies >f.sub.N) the lower sideband U' is present, on the
contrary, at the two outputs of the mixer stages in the same phase
position, whereas the upper sideband O' is in the opposite phase position.
This situation is also shown in FIG. 2d and FIG. 2e.
The different phase positions of the sidebands U and O in the lower
frequency band can be used for suppressing one of the sidebands. To this
end an adder (subtractor) circuit 10 is provided to whose inputs the
output signals u.sub.2 and u.sub.2 ' of the two mixing stages 3a and 3b
are supplied and which is preferably so constructed that the two signals
may optionally either be added to or be substracted from one another, so
that either the lower or the upper sideband is suppressed in the lower
frequency band. In the upper frequency band the other sideband is then
suppressed. FIG. 2f shows the output signal u.sub.3 of the circuit 10
after adding (subtracting). This output signal is applied to a low-pass
filter 11, which suppresses all frequencies above the oscillator frequency
f.sub.N, as illustrated by the dashed representation of the upper
frequency band. Consequently, only the original upper or the lower
sideband--depending on whether the output signals of the mixing stages 3a
and 3b are added or subtracted in the circuit 10, remains at the output of
the low-pass filter. Unfortunately, the low frequency information is in
the inverted frequency position so that a further mixing operation is
required in mixer 13 for obtaining the correct position.
The above explanations will make it clear that the suppression of one
sideband depends on as accurate as possible a compensation in the circuit
10 of the two components of the output signals of the mixer stages 3a and
3b corresponding to this sideband and on that the upper frequency band is
suppressed as far as possible by the low-pas filter 11.
It is a requirement for the accurate compensation of either of the two
sidebands that the two branches comprising the elements 1a, 2a, 3a, 15, 7a
and respectively 1b, 2b, 3b, 16, 7b are as identical as possible. The best
possible way to achieve this is by combining corresponding components in
an integrated circuit.
To obtain a suppression of the upper frequency band which is as accurate as
possible the low-pass filter 11 is preferably a gyrator--low-pass filter,
which is constructed so, that its first pole position is substantially
equal to the frequency of the oscillator 8. To this end a control circuit
can be formed, comprising a phase comparison circuit 12 producing a signal
which depends on the phase difference between the voltages at the input
and the output of the low-pass filter 11, which signal is utilized for
adjustment. Use is made of the fact that in the vicinity of the pole
position of the attenuating filter this phase difference is highly
dependent on the frequency. Instead of controlling the pole position of
the low pass filter 11, it is also possible to control the frequency of
the oscillator 8 by means of the phase comparison circuit 12. This is
indicated in FIG. 1a.
The output signal of the low-pass filter 11 comprises the upper or the
lower sideband in the inverted frequency position (inverted LF-band), that
is to say the higher frequency components of the sideband correspond to
lower useful frequencies and the lower frequencies in the sideband
correspond to higher useful frequencies. To bring the given sideband in
the correct receiving position, a last mixing stage 13 is provided in
which the output signal of the low-pass filter is mixed with the signal
produced by oscillator 8. Last mentioned stage 13 must be--like the mixing
stages 3a and 3b--a multiplicative mixing stage, for example a product
detector, so that its output signal contains only the sum and difference
frequencies formed from its two input signals (cf. FIG. 2g). The sum
frequencies then produced, which represent the given sideband U" or O" in
the inverted position, are removed by means of a low-pass filter 14, whose
cutoff frequency also corresponds to the frequency f.sub.N of oscillator
8. The output signal of last-mentioned low-pass filter represents the
useful low-frequency signal.
As mentioned above, the effects of the selective fading are avoided in the
circuit arrangement according to the invention by using one of the two
sidebands U or O for reception. The adder (subtractor) circuit 10 can then
be selected to operate in the mode (adding or subtracting) in which the
less disturbed sideband is received. As only one sideband is used for
reception, two sidebands which are independent from one another can also
be received with this circuit arrangement.
As the frequency of oscillator 4 corresponds with exactly the same phase to
the carrier frequency, it is impossible that, owing to the radiation of
this frequency, interferences are produced, neither in the receiver
comprising the circuit according to the invention, nor in the receivers. A
remainder of the frequency f.sub.N which may still be present in the
output signal of the mixing stage 13 can be suppressed without further
measures by the low-pass filter 14. Coils are not required in the overall
receiver so that the circuit can be implemented without further measures
in integrated circuit technology. It is not required then to combine all
the modules in one single integrated circuit, but it is important that
modules which must have the same characteristic, for example the mixing
stages 3a and 3b, are integrated on one and the same semiconductor
substrate. In principle, it is alternatively possible to receive double
sideband signals without carrier with the circuit according to the
invention. In this case, however, a PLL-circuit cannot be used and the
frequency of the oscillator 4 must very accurately correspond to the
frequency of the suppressed carrier. In the most simple case the usual
variably tuned input selectivity can be fully dispensed with, or it is
possible to use input band filters which are fixedly adjusted to the
receiving band.
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