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BACKGROUND OF INVENTION
Low voltage d.c. power sources are widely used in manufactures of many
different kinds. Most known low voltage d.c. power supplies, when operated
from a source of, for example, utility line voltage (e.g., 117 volts, 220
volts, etc.) employ the well known step-down transformer, rectifier, and
usually feedback regulator circuit means so as to provide a d.c. voltage
having good constancy and relative freedom from line voltage excursions.
Such means are oftimes expensive and complex when compared to the product
in which they are intended to find application. A real need exists, which
my invention serves to satisfy, for an inexpensive, small, easy to
mass-produce low d.c. voltage power supply which has good stability
characteristics and high efficiency. In particular, the desirability for
such a d.c. power supply is ever-increasing as solid-state controls, for
example microprocessors or the like, reach into lower priced mass-produced
consumer products.
In an earlier U.S. Pat. No. 3,355,650 Tolmie describes an arrangement for
delivering a nominal d.c. value to a motor e.g., electric shaver or the
like) from an a.c. mains which may vary. However, the inherent stability
of the so-produced d.c. voltage is poor relative to what is required to
satisfy more demanding electric loads, such as presented by control
circuits including a microprocessor or the equivalent. This instability is
described in the "SCR Manual Including Triacs and Other Thyristors", fifth
edition 1972 publication ETRC-3875B by Semiconductor Products Department,
General Electric Co., Syracuse, N.Y. 13201. Chapter 4.10, page 93
paragraph 2 particularly describes the phenomenon: "The transfer function
is very non-linear and repeatibility of setting is not possible either
with different SCR's or with temperature due to I.sub.GT variation." The
"transfer function" describes the point where the SCR turns "on" when the
only source of gate current is a substantial resistance coupled to the
anode thereof. The class of triggering described therein is also that
taught by Tolmie. In addition, the arrangement brings about a premature
device failure mode whenever the SCR current rating is cost effectively
near the a.c. load demand. The cause is well described in the "Thyristors,
Rectifiers, and Diacs" databook SSD-206A 1973 edition published by R.C.A.
Solid State Division, Box 3200, Somerville, N.J. 08876. Reference to page
438, paragraph 2 which effectively says that: " . . . it is always
advantageous to provide a gate current pulse that has a magnitude
exceeding the d.c. value required to trigger the device." While on page
442, paragraph 1 states: "Although the circuit is capable of providing
variable power to the load, it is heavily dependent on the gate current
distribution, and results in uncontrolled conduction angles for a given
value of gate series resistance." The instant invention overcomes these
earlier difficulties, through the novel inclusion of an abrupt breakdown
voltage responsive means coupled directly between the anode and the gate
of the s.c.r., or its equivalent, so as to provide rapid infusion of gate
current once the threshold of the voltage responsive means is reached.
This action provides a highly stable unipolar interrupter signal which is
filtered as a source of constant d.c. potential, while at the same time
serves to deliver the remaining part of the a.c. power cycle to the a.c.
load work function without any important loss of effectiveness.
My invention's teachings derive the low d.c. voltage directly from the
primary power line, thus there is no inherent isolation as might be
provided by oridinary power supplies using transformers. The invention
proposes, however, that in cost-effective products this presents no
particular technical problem and any disadvantages are offset by cost
savings, as line voltage operated relay controllers, timers, and the like
have been used for years without difficulty. Furthermore radio,
television, phonograph and other consumer oriented products are known to
have employed direct connection between the a.c. primary line and the
internal workings.
In the most basic expression of my invention merely five circuit elements,
employed in novel arrangement, serve to acheive performance results which
prior to my finding would require substantially more complicated and
costly apparatus.
SUMMARY OF INVENTION
A power supply is described which provides a stable source of low direct
current voltage by way of utilizing a small portion of the alternating
current cycle of an a.c. load terminated a.c. source. Operation is
dependent upon the action of a variable impedance means or the like
coupled between the a.c. source and the a.c. load, and the resultant
effect of interrupting the current flow between the a.c. source and the
a.c. load for a small percentage of at least one half of each full a.c.
cycle, thereby diverting the voltage developed across the variable
impedance interrupting mechanisim through a secondary current path means
which serves to charge an energy storage capacitor to an average d.c.
value.
A high speed switch, such as a thyristor, is used as the interruptor
mechanism. The switch will repeatedly commutate, or effectively open, when
the alternating current potential across the switch is near zero value,
whereupon through at least some part of one half of the full a.c. cycle it
will remain effectively open, or at a higher impedance value, until the
a.c. peak voltage developed across the switch reaches a finite magnitude,
whereupon the switch will turn-on, or close, thereby appearing as a low
impedance which serves to decrease the drop across the switch to a minimum
value. During the brief period occurring between zero a.c. value and the
finite value where switch-over occurs from the higher to the lower
impedance value, a substantially unidirectional potential is developed
across the switch means which serves to pulse charge a storage, viz
filter, capacitor, the combination thereby producing a source of d.c.
potential exhibiting good constancy.
Therefore the prime object of my instant invention is to teach a d.c. power
supply which develops its value by way of diverting but a small portion of
the power intrinsic in at least a part of one half of the full a.c. cycle
in a current path between a source of a.c. power and an a.c. power load.
Another important object of my invention is to provide a d.c. power supply
which operates off of but a small portion of the current flowing in an
a.c. power circuit with the value of the developed d.c. level being
substantially independent of nominal variation in either current or
voltage in the a.c. source.
Another significant object of my invention is to provide a d.c. power
supply which requires no magnetic components, e.g. transformers or the
like, in its implementation and yet is capable of providing relatively low
d.c. voltage values having good constancy from a source of relatively high
voltage a.c. power having poor regulation.
Still another object of my invention is to develop a low d.c. value having
good regulation from a source of poorly regulated high a.c. value whilst
maintaining good conversion efficiency, compact size, and minimum circuit
cost.
Yet another object of my invention is to give rise to a means which will
develop more than one relatively stable low voltage d.c. value from a
common high voltage a.c. source and load circuit means and having a wide
latitude for any combination of voltage, current, and frequency
instability in the a.c. source and load circuit.
A remaining object of my invention is to teach the use of a thyristor as a
switch means substantially arranged in series connection between an a.c.
source and an a.c. load, whereby the thyristor appears as an open circuit,
or high impedance, through at least a few electrical degrees of at least
one half of each a.c. full cycle, whereafter the thyristor is turned on,
or effective as a low impedance, by a voltage responsive means, as for
example a zener diode, throughout the balance of the interrupted a.c. half
cycle resulting in good a.c. load continuity with the a.c. source while
yet providing at least one unilateral potential during the open circuit
period, as developed across the thyristor, which serves to charge an
attendent energy storage capacitor the result of which is to provide a
secondary circuit d.c. source of power.
BRIEF DESCRIPTION OF DRAWINGS
FIG. 1 The use of an unilateral thyristor switch element is shown together
with a separate alternate half-cycle bypass diode.
FIG. 2 The use of a bilateral thyristor providing full first half-cycle
a.c. power conduction and controlled second half-cycle a.c. power
conduction.
FIG. 3 A bilateral thyristor provides double controlled alternate
half-cycle a.c. power conduction and develops at least two d.c. output
levels.
FIG. 4 A double controlled bilateral thryistor combined with a bridge
rectifier provides double recurrence charging of the d.c. energy storage
capacitor.
FIG. 5 The position of the voltage level sensing breakdown diode is shown
changed over the arrangement of FIG. 1 to a position directly supplied by
the average d.c. output voltage.
FIG. 6 The use of a transistor gain element interconnected between the
breakdown diode and the thyristor gate is shown.
FIG. 7 A voltage comparator serves to provide a d.c. voltage output
relatively independent of the absolute value of the breakdown diode.
FIG. 8 Plot showing voltage regulation performance with change in current
load for the circuit described in FIG. 1.
FIG. 9 Plot showing voltage regulation performance with change in current
load for the circuit described in FIG. 2.
FIG. 10 Plot showing the tracking and voltage regulation performance with
change in current load for the dual output circuit described in FIG. 3.
FIG. 11 Plot showing output voltage regulation with change in the input
a.c. voltage value for circuit described in FIG. 1.
FIG. 12 Plot showing output voltage regulation with change in the input
a.c. voltage value for circuit described in FIG. 6.
FIG. 13 The series arrangement of two power supply elements is shown in an
arrangement which will result in a multiplicity of d.c. output values.
FIG. 14 Plot showing voltage regulation with change in load current for a
particular embodiment of the circuit described in FIG. 13.
DESCRIPTION OF INVENTION
A load means 1 is shown in FIG. 1 to connect on one end to primary power
line L1, while the other end connects in the main through thyristor means
101 to primary power line L2. With a primary power source of alternating
current, for example a source of typical 60 hertz (50 hertz) utility
power, connected between L1 and L2, circuit operation is believed to
behave as herein described. On the half-cycle when L2 is positive with
respect to L1, power will pass directly to the load except for a minor
junction drop across the diode 104, that being on the order of 0.7 volts
with the usual diode. However on the alternate half-cycle when input line
L2 swings negative with respect to line L1, the diode 101 will not
conduct. Furthermore, the thyristor 101 which acts as a circuit
interrupter will not conduct until the gate of the thyristor, which is
held near the cathode potential by anti-leakage swamping resistor 103,
receives ample current to initiate turn-on. What happens is as the primary
voltage increases in value from zero, the voltage on the anode of the
thyristor 101 will proportionately increase as a a positive value with
respect to the cathode.
This increase in voltage, or interrupter signal, continues until the zener
breakdown diode 102 conducts, producing an abrupt increase of current flow
in the gate circuit of the interrupter means thyristor 101. This causes
the thyristor to conduct, and the primary line L2 will connect directly to
the load throughout the rest of the power half-cycle.
With a typical zener diode 102 value of 6.8 volts, by way of example, the
interrupter signal voltage across the thyristor will increase at least to:
E.sub.1 =V.sub.z +E.sub.GK
where:
E.sub.1 =voltage across thyristor at turn-on
V.sub.z =zener breakdown voltage
E.sub.GK =thyristor gate to cathode voltage at turn-on
prior to turn on state; therefore for the cited example, where V.sub.z =6.8
volts and E.sub.GK may be 0.7 volt, E.sub.1 will equal about 7.5 volts.
This interrupter signal voltage E.sub.1 couples through diode 200 to a
charge holding, viz filter capacitor 201 and serves as a signal adaptive
means. The result is that a stable d.c. voltage level, the magnitude of
which is near the value V.sub.z, will be developed across the capacitor
201, thereby serving to energize load means 2. As shown, a voltmeter 25
may be connected across the load 2 and read the developed d.c. value.
The practitioner of my art will recognize that, with typical a.c. primary
line voltage 117 volts (root mean square) the peak voltage will
approximate 165 volts; therefore with the typical zener value of 6.8 volts
as described, the ratio relative to the line voltage is small, meaning the
percentage of d.c. imbalance or power loss presented to the load will be
negligible in the majority of practical applications for my teachings,
being in the cited example only around 4% of the half cycle value.
Furthermore, with the example given in a circuit wherein the filter
capacitor 201 is on the order of 150 microfarads, an available current of
25 milliamperes or more at a d.c. potential around 6 volts may be
expected.
A bilateral thyristor, or Triac, is shown in FIG. 2. The thyristor 102
connects between the input line L2 and the load 1 and functions as an
interrupter means. On the power half-cycle when L2 is positive, the gate
of the thyristor 102 connects directly to the main terminal two (MT-2) as
connected to the load, by diode 110 which is forward conducting through
peak current limiting resistor 111, say 35 ohms, thereby causing the
thyristor 102 to conduct essentially over the full half-cycle. Resistor
112 serves as a gate leakage swamping resistor, being on the order of 470
ohms.
When input L2 receives the alternate negative half-cycle, diode 110
operates as a breakdown, or zener avalanche, diode thereby effecting a
belated turnon of the thyristor until the voltage increases across the
thyristor to the breakdown value V.sub.z for the selected zener diode,
whereupon current is abruptly coupled to the thyristor gate electrode. For
example, with a 1N753 diode, a value of about 6.2 volts, plus the V.sub.GT
(e.g., the gate to MT-1 drop) is required. This interrupter signal voltage
is coupled to the signal adaptive means including diode 200' coupled to
energy storage, or filter capacitor 201 and is available for supplying the
load 2. With a filter capacitor of 270 microfarads, and using a 1N753
zener diode as described, the supply has satisfactorily supplied 5.8 volts
at a load of 25 milliamperes.
A dual polarity voltage power supply means is shown in FIG. 3. A bilateral
thyristor 102 is caused to have retarded turn-on by way of the
back-to-back zener diode arrangement 120, 121. As shown, the resulting
d.c. voltage positive value will be for the most part determined by the
avalanche characteristics of zener diode 120 whilst the negative value
will be mainly set by the breakdown value of zener diode 121.
When MT-2 of thyristor 102 goes positive to the value pre-established by
zener 120, the d.c. value will be coupled by diode 210 to filter capacitor
211. Conversely, when MT-2 of thyristor 102 goes negative to the value
pre-established by zener 121, the d.c. value will be coupled by diode 215
to filter capacitor 216.
The resulting positive value +V.sub.c and negative value -V.sub.c developed
across the filter capacitors couples to the respective d.c. loads 2A, 2B.
From this teaching an artisian will quickly conclude that a variety of
positive and negative d.c. levels may be obtained, since the individual
value is determined by the intrinsic voltage breakdown value of each of
the zener diodes 120, 121 and may be purposefully different.
Full-wave utilization of the primary power is depicted in FIG. 5. In the
previous cited examples relative to FIG. 1 and FIG. 2, half-wave primary
power utilization is employed, with the result that considerable time
(e.g., 16.67 mS for 60 hertz; 20 mS for 50 hertz) elapses between charge
pulses for the filter capacitor. The result is the capacitor has to be
larger and more costly to acheive a particular level of performance.
Through the effective expedient of employing four low voltage diodes, only
three more than otherwise needed in FIG. 2; together with an additional
zener diode as in FIG. 3, considerable improvement in regulation and
ripple reduction may be obtained. The four diodes 220, 221, 222, 223
connect as a bridge rectifier between the thyristor 102 and the filter
capacitor 225, therefrom to the load 2.
The output d.c. voltage is about the sum of the individual zener 120, 121
values, expressed as:
V.sub.c .perspectiveto.E.sub.1 +E.sub.2 =(V.sub.Z1 +V.sub.Z2)+2E.sub.T
where:
E.sub.1 =voltage across thyristor at turn-on for first half cycle;
E.sub.2 =voltage across thyristor at turn-on for second half cycle;
V.sub.Z1 =Zener 120 breakdown voltage;
V.sub.Z2 =Zener 121 breakdown voltage; and,
E.sub.T =thyristor gate trigger voltage to MT-1.
The avalanche diode 102' is connected between the d.c. value coupled from
the thyristor 101 anode by diode 200 and the thyristor gate in the
depiction of FIG. 5. The result is the gate-firing of the thyristor is
determined by the maximum d.c. level, not the instantaneous peak anode
voltage. The effect is some reduction in the dynamic impedance of the
power supply thereby effecting an improvement in regulation.
A dynamic, viz amplified, control means appears in FIG. 6. A PNP transistor
240 (2N6076 for example) is connected with the emitter to the +V.sub.c
line and the collector to the thyristor 101 gate. When the +V.sub.c d.c.
value across the filter capacitor 201 is low, e.g. less than the zener 241
breakdown value, the transistor 240 base is pulled to the emitter by
resistor 242 (say 680 ohms) resulting in no collector current flow, the
ensuring effect of which is to cause the gate of the thyristor 101 to be
pulled to the cathode potential by way of resistor 243 (say 3,900 ohms).
The thyristor will not conduct, producing a positive voltage transfer
through diode 200 which, a knowledgeable artisian will understand, serves
to charge capacitor 201. When the charge on the capacitor 201 rises to a
value where the avalanche diode 241 reaches zener breakdown, base current
will be caused to flow in transistor 240, the intrinsic gain of which will
produce substantial collector current, thereby producing an abrupt gate
current increase in the thyristor 101, subsequently causing it to conduct,
or turn-on, and thereby inhibiting any further charging of capacitor 201.
The gain of an operational amplifier 230, connected as a voltage
comparator, provides enhanced performance of the circuit depicted in FIG.
7. As shown the thyristor 101, breakdown diode 102, swamping resistor 103
and alternate half-cycle pass diode 104 function as described for FIG. 1.
except that the breakdown diode 102 is selected to have a value
substantially higher than the normal operating d.c. output value expected
from the circuit. The breakdown diode 102 functions, in this arrangement,
as an over-voltage protection diode and, therefore, may be deleted in some
applications to save cost. Diode 102 may for example have a zener value of
say 15 volts. The constancy of the actual d.c. output voltage is
determined principally by zener diode 231, as connected to the inverting
input of operational amplifier 230 (say type CA-3140) along with zener
ballast resistor 232. The amplifier 230 non-inverting input, e.g. "+"
input, serves to connect to a voltage divider including resistors 235, 236
and usually potentiometer 234. The actual d.c. voltage provided by the
power supply is determined by the ratio of the resistance R.sub.M, being
that between the potentiometer 234 arm and +V.sub.c including resistor
235; and resistance R.sub.N, being that between the potentiometer 234 arm
and -V.sub.c, including resistor 236; shown as:
##EQU1##
where: V.sub.DC =+V.sub.c output level; and
V.sub.ZR =Avalanche voltage for zener reference diode 231.
As long as +V.sub.c as ratioed, e.g. divided, and connected to input 237 of
amplifier 230 is of a value LOWER than the zener reference value on input
233, the amplifier output will be minimum, near zero or -V.sub.c. The
result is no substantial current will conduct through steering diode 238
into the gate of the thyristor 101, until the aforesaid zener 102
conducts. The resultant positive value developed across the thyristor 101
will couple through diode 200", serving to charge capacitor 225. Depending
on the value of the capacitor 225, one or more a.c. cycles may occur
before the d.c. value thereon accumulated rises sufficiently to cause the
positive value on amplifier 230 input 237 to exceed than on input 233.
When this does occur, the amplifier 230 output will drive to near
+V.sub.c, thereby turning "on" the thyristor and stopping further
capacitor 225 charging through the rest of any particular a.c. cycle.
A resistor 130 serves to current limit the d.c. output and can best be
understood if one considers that all of the d.c. current passes through
the resistor. The drop produced in the resistor, when it is of low value
say 0.47 ohm, will be negligible for normal power supply output currents
typified by 25 to 50 mA or so. However if a fault current occurs (shorted
output, etc.) the drop across the resistor will increase until the gate of
the thyristor 101 is effectively positive by the path through resistor
103. Thus the thyristor will turn-on, negating any further current flow
during the normal charge half-cycle. The action is repetative for every
half-cycle until the fault is removed, thus providing effective current
fault limiting.
The performance curve for a particular embodiment of my invention appears
in FIG. 8. What is shown is the change in +V.sub.c, or d.c. output volts,
with any change in d.c. load current for the circuit arrangement described
in FIG. 1. Two curves appear on the plot: that curve including points AA,
AB wherein the value of capacitor 201 is 270 microfarads (uFD); and that
curve including points AC, AD wherein the value of capacitor 201 is 820
uFD. The zener diode 102 is a 1N753 (6.2 V.sub.z) and the thyristor is a
C106B. The range between points AA, AB and AC, AD represents a 2:1 change
in d.c. current. With the example between points AC, AD the measurement
shows only a change from 5.45 to 5.3 d.c. volts as current doubles from 20
to 40 mA d.c. This is merely a 2.75% change in voltage for a 2:1 current
change: thus the essence of the invention is illustrated, that being a
voltage stable power supply comprising a modest arrangement of inexpensive
elements.
In a like way FIG. 9 shows the measured performance for a particular
embodiment of the circuit described in FIG. 5. As before, the range
between points BA, BB and BC, BD bounds a 2:1 current change and
demonstrates what good regulation is achievable. In this configuration,
capacitor 201 is 200 uFD for curve BA, BB and 500 uFD for curve BC, BD.
The zener diode 102 is a 1N753.
The teaching of the circuit for FIG. 6 is shown in the curves appearing in
FIG. 10. As before, the two curves encompass points CA, CB for a capacitor
201 value of 200 uFD, and points CC, CD for a capacitor 201 value of 500
uFD.
Line voltage effects on the d.c. output of the circuit shown in FIG. 6
appears in the curve of FIG. 11. The value for capacitor 201 is 500 uFD
with a load current of 20 mA. The point DA represents the usual 95 volt
accepted "minimum" line voltage which may be expected from the typical 117
volt a.c. source. The a.c. load is a 100 watt incandescent lamp bulb. One
novel improvement the artisian will realize from my teachings is that
higher line voltage, e.g. 230 volts or more, have no material effect on
the performance of my device. Furthermore, in the circuit as in FIG. 1,
the peak reverse voltage V.sub.ROM across the thyristor is negligible,
being on the order of the forward junction drop of diode 104, whilst the
diode reverse voltage V.sub.RM is on the order of the value of the
breakdown potential of zener 102. Thus it becomes obvious from my
teachings that lowest cost, low peak-reverse voltage semiconductors
perform well in the circuit.
FIG. 12 depicts the measured performance for the circuit of FIG. 5 with a
C201 value of 200 uFD and a load current of 40 mA d.c. The a.c. load is a
100 watt lamp. What is shown in each FIG. 11 and FIG. 12 is the
extraordinary insensitivity to line voltage variation inherent in my new
invention.
A multiple value d.c. output power supply, that is one providing more d.c.
level values than taught by FIG. 3, is shown in FIG. 13. In effect what I
give the art as novel in this teaching is the series arrangement, or
stacking, of two or more of my aforesaid power supply forms. The artisian
will quickly realize that the depicted power supply comprises two earlier
described power supply variants, that of FIG. 1 and that of FIG. 3. The
result is that in addition to the negative -V.sub.c and positive +V.sub.ca
value shown as to be provided in a method similar to the like outputs for
earlier FIG. 3, this supply provides a second positive +V.sub.cb value
somewhat higher in voltage than +V.sub.ca. What occurs in the interrupter
signal effected by the thyristor (triac) 102 is rectified by either diode
200A for +V.sub.ca, or else diode 215 for -V.sub.c. At the same time, the
interrupter signal effected by thyristor (s.c.r.) 101 is rectified by
diode 200B so as to develop voltage +V.sub.cb. Since thyristor 101 is
effectively in series with thyristor 102, MT-1 of device 102 being
"common", three distinct voltage values are produced.
FIG. 14 provides a plot of a particular embodiment of the multiple value
power supply which finds application for providing power to an integrated
circuit device such as the ubiquitous type 8080A microprocessor, for
example. Zener diode 102 is exampled as a 1N756 (8.2 V.sub.z), whilst
diodes 120, 121 are each type 1N752 (5.6 V.sub.z). Each output capacitor
201', 211, and 216 is 1 millifarad. The curve F shows three distinctive
plots for +V.sub.ca, +V.sub.cb, and -V.sub.c. The points between FA, FB;
FC, FD; and FE, FF on each plot represents a 2:1 current change, giving a
showing of how very little current changes affect the output voltage.
While my teachings give example of particular combinations of active and
passive components and values, the practitioner will understand that any
of a very large combination of element values may be employed with similar
results and yet not deviate from the essence of my invention.
My teachings provide a d.c. power supply capable of providing low d.c.
voltages in the range of around 1 volt to about 15 volts d.c., more or
less, with a current capability from near zero to around 100 or more
milliamperes d.c. The exact values for circuit operation are best selected
by the artisian with consideration as to the character of the cooperative
d.c. load means, and therefore, such values do not provide substance as to
establishing limits on the scope of my instant invention.
The term a.c. source shall include source means of any frequency
what-so-ever within the operating limits of the thyristor selected for the
task at hand, e.g. 50 hertz, 60 hertz, 400 hertz, etc.
The term a.c. source further defines a waveform envelope which has no
significant d.c. component, viz goes through a polarity reversal during a
substantial part of the cycle period. Commercial utility power is a good
example.
While it will be obvious to an artisian in the field, I will specify that
the a.c. source waveform shall have a rate of rise (e.g., leading edge
rise time) which is a substantial portion of 90 electrical degrees of the
operative a.c. half-cycle. This is to say, the waveform is best as a
sinusoidal, triangular, or sawtooth waveshape. A good squarewave, having
fast rise times, such as generated by certain kinds of inverter circuits
and by particular classes of constant voltage regulators will require
waveform shaping by way of a filter or the like in order to acheive best
performance from my teachings.
The cascade connection, viz series arrangement, of two or more thyristors
is contemplated as being a method for developing a plurality of more than
two distinctly different output d.c. values. Therefore the implementation
of such an arrangement to develop, say -5, +5, and +12 volts d.c., would
be ordinary practice of my new teachings.
The compound connexion of two unilateral acting thyristors, e.g. the
inverse parallel connective arrangement, for the purpose to produce more
than one d.c. output value is considered an expediential application for
the teachings of FIG. 1. This particular embodiment variation includes
that arrangement wherein the cathode of one thyristor couples to the anode
of the other and each thyristor is acted upon in a substantially separate
way by a zener diode or like control means effectively coupled
cooperatively between the anode and the gate of each thyristor, thereby
usually producing a separate interrupter signal value for each half of the
a.c. cycle.
The arrangements taught in my instant invention have been modelled as
encapsulated forms, with the interconnective circuitry being a printed
circuit board. The combination has then been impregnated with a plastic
substance to effect a module assembly about 40 millimeters long, 30
millimeters wide, and only about 12 millimeters thick, with leads
extending therefrom. As such the module weight is about one ounce.
* * * * *
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