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Description  |
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FIELD OF INVENTION
My invention relates to fractional and integral horsepower electric
induction motors particularly of the kind manufactured in large volume for
use on major appliances, refrigerators, and air conditioners which drive a
load that routinely varies over a range between full load and less than
full load. The principal object for my invention is to teach a controller
which serves to produce considerable ENERGY SAVINGS through load-related
reduction of power flow to the motor under all but the full load
conditions.
BACKGROUND OF INVENTION
A.c. induction motors find ubiquitous application in major appliances,
refrigerators, air-conditioners, and other machines of all sorts.
Induction motors are cheap and simple to manufacture, and have an enviable
record for long-term reliability without attention. Induction motors are
relatively efficient electrically, when fully loaded. When lightly loaded,
they also are notorious for wasting considerable amounts of electricity by
consuming far more electrical power than what they are called upon to
deliver as operating torque from their output shaft. It is this later
rather troublesome shortcoming of common induction motors which needs
improvement, and it is believed that this invention now can offer
considerable relief.
To give some scope to the impact which induction motors have on society,
one may consider that "125 million household refrigerators and freezers in
operation today require the electricity from 30 standard (large sized)
1,000-megawatt power plants." ("Scientific American", Vol. 258, No. 4,
April 1988, Page 56, in an article `Energy-efficient Buildings` by Arthur
Rosenfeld and David Hafemeister.) Virtually all such refrigerators use
induction motors. Considerable waste occurs because refrigerators do
operate with varying mechanical load demands, while the typical
hermetically sealed compressor assembly contains an induction motor which
is sized to handle the worst case, ableit `normal`, compression load
demand imposed upon it without frequent stalling. The result is simple:
much of the time the motor is operating at less than full load and wasting
a considerable amount of energy. Merely improving the dynamic operating
efficiency of such a motor a mere ten-percent or so may result in the
power saving equivalent to that afforded by 3 of the large 1,000 megawatt
power plants, which are said to cost several billion dollars to construct.
Some perspective of what this means can be obtained by considering an
article which appeared in "New England Business" magazine, May 2, 1988, on
pages 39-40 wherein the New England Power Pool (an organization which
represents the region's utilities) estimates that by 1995 about
4,000-megawatts of additional generating capacity (on top of the present
23,000-megawatts of current capacity) would be needed just to keep up with
demand growth. You also need to keep in mind that the highly controversial
New Hampshire `Seabrook Unit I` and Plymouth, Mass. `Pilgram` nuclear
power generating facitlities produce a combined power of only
1,820-megawatts: far less than what might be conserved through better
induction motor operating efficiencies!Hence, improved efficiency in motor
operation for refrigerators, air-conditioners, and other machines could
down-scale the demand growth and alleviate some of the pressures which now
exist in getting additional capacity on-line. Needless to say, greater
improvements in motor efficiency can afford even more spectacular economic
savings in power plant needs and reductions in attendant `wasteful`
consumption of non-renewable fuel resources. Such further improvement in
induction motor operating efficiency is precisely what is brought about by
my instant invention.
Modern induction motors are often designed to operate with very high
magnetic field flux densities in the stator structure. The result is
near-saturation of the core material, with high eddy current losses. In
addition, the windings may be designed to operate with high current
densities that results in considerable heating due to winding resistance
losses. Such winding losses are further aggravated in many cheaply
designed appliance motors through the use of aluminum wire in lieu of the
better and generally more efficient copper wire windings. Motor design my
be dictated by competitive market conditions, resulting in agressive cost
cutting. Cheap designs commonly translate into producing motors having
high operating levels and the result may be a motor which operates with
reasonable efficiency under full load, while under light load it is a
wasteful of considerable energy. High temperature rise in a lightly loaded
(or unloaded) motor is a sure sign indicating poor electrical operating
efficiency. Modern motors operate very hot under all conditions of
loading, which translates into poor overall performance efficiency when a
widely varying load is being driven by the motor.
In earlier U.S. Pat. Nos. 4,052,648 and 4,266,177 Frank Nola describes how
the a.c. pwoer fed to an induction motor might be controlled and therefore
bring about some improvement in electrical efficiency. While he does
measure the power factor of the operating motor and therefrom determines
various control values for regulating the total power input of the motor
run winding set by conventional phase-angle controlled firing of a triac,
he greatly reduces and in some cases may negate any purported improvement
by virtue of the severe a.c. power waveform harmonic distortion which his
system reflects into the electric utility system. Nola also describes
apparatus which is fraught with possibilities for maladjustment and drift,
and wherein the correct operating points are not pre-established by design
but rather they must be somewhat tailored to each unit which might be
manufactured, resulting in a relatively labor-intensive and costly
product. Column 3, lines 40-47 and column 6, lines 50-66 of U.S. Pat. No.
4,266,177 particularly describes the kind of twiddling that is needed to
set the device's operating points relative with any particular motor's
observed performance.
In yet another U.S. Pat. No. 4,533,857, Ten-Ho Chang et al show a motor
controller which in effect measures the apparent current drawn by an
induction motor and therefrom develops a variously retarded phase-angle
control signal which fires a triac and thus modulates the total power flow
to the motor. Like Nola, Chang et al shows the turn-ON of the full motor
running current at some delayed point during each a.c. half-cycle and of
course such an approach is fraught with severe harmonic distortion of the
a.c. power flow (as reflected into the a.c. power lines), accompanied by
resulting inefficiencies that may exceed any gain which could otherwise be
obtained from use of the controller. Chang also does not recognize nor
allow for the common characteristic of cheap induction motors wherein the
lightly loaded (or even unloaded) apparent motor current may be only a
little less than what full load motor current is. Although the motor
current is lagging by perhaps 60 degrees or more, the actual measurable
current which develops across the current transformer (Tr-2 in Chang's
teaching) will be quite nearly the same as what develops under full motor
load, when the motor current might lag by 30-degrees or less. A typical
appliance motor, such as the General Electric type 5KH46JR15S has been
found to draw about 7.9 amperes under full load, and yet continue to draw
nearly 7 amperes of apparent current under NO-LOAD. Power factor also
varied from about 80-85% under FULL-LOAD, to about 20-30% under NO-LOAD.
This of course means that little change in current occurs and the circuit
of Chang would operate ineffectively because slight changes in a.c. line
voltage bring about more substantial changes in motor current than what
changes in motor load produce. Chang is silent regarding compensation of
apparent motor current changes which merely relate to commonplace a.c.
line voltage fluctuations.
Noise, in the form of hum and buzz, are byproducts of stressful motor
operating conditions. Magnetostrictive effects tend to produce substantial
noise in the motor's structure, paticularly when stressed with the strong
and abrupt changes in flux brought about by phase-delayed thyristor power
control. These abrupt changes in induction fields can also set up other
parasitic vibrations which, aside from being audibly annoying, can lead to
premature structural fatigue of the motor's components (such as a
vibrating loop of wire which eventually breaks off). Refrigerators and, to
a lesser extent, air conditioners are frequently annoying sources of
audible noise, albeit not particularly high in the sense of loudness on
the commonly cited decibel scale for noise sources. Load related
modulation of power flow to such motors may therefore serve to
substantially abate such undesirable noise and result in a more acceptable
product.
SUMMARY
A.c. induction motors provide a very economical and time proven source of
mechanical power for driving major appliances, air conditioners, and other
kinds of domestic and commercial machines. The time-proven dependability
of induction motors is exceptional, and years of product engineering have,
in most cases, resulted in a simple and cost effective configuration using
few parts. It is thus desirable to retain all these advantages of the
induction motor, while coming forth with a meaningful reduction in energy
waste which occurs when the motor is less than fully loaded.
Production of electric power in America is reaching a point where the
utility companies in many parts of the nation will soon be nearing 100%
capacity. Unless more generating capacity is soon built, brown-outs, power
grid failures, and other cataclysmic power distribution events are likely
to occur with increasing frequency because no reserve power capacity is
available or being readied. In view of the many years it takes to get even
a single new nuclear or conventional electric power generating facility
on-line means that there is no quick and simple solution to this looming
energy-crunch dilemma. The building of additional power plants is also a
fundamentally costly proposition. Such cost can be illustrated by a
500-megawatt gas-fired power plant located in Burrillville, R.I. which
cost about $320-million dollars and by a $300-million dollar plant planned
for Woonsocket, R.I. which is oil-fired and produces a mere 180-megawatts.
It therefore behooves manufacturers of any kind of apparatus that tends to
waste a lot of electricity, relative to what it really "uses" to drive a
load, to develop more ENERGY EFFICIENT schemes. Paramount in this arena of
everyday power wasters are the ubiquitous induction motors, such as found
on most major appliances and in air conditioners. Induction motors are
subject for being `singled out` as power wasters due to the hunge number
of such motors which find extended operation every day in out modern
society. They often power machines and appliances which regularly operate
daily for substantial periods of time. It is common that induction motors
are desired by appliance and machine designers for any application where
the motor will see a lot of use, due to their time-proven reliability and
nearly zero-maintenance requirements. They also lend themselves to
hermetic refrigeration compressor assemblies because there are no brushes
to wear out or cause contamination of the refrigerant (and oil).
Ordinary engineering practice produces induction motor designs which
operate with high magnetic field flux densities, high current density
through the windings, and with a minimum of structure. The General
Electric `Form V` industry standard no. 4096 motor, typically used with
Whirlpool and Kenmore brand electric clothes dryers is representative of
such minimal modern design practice. Producing about 1/3 horsepower, this
motor draws about 5.6 amperes (full load), while under reduced load the
apparent motor current remains about 5 amperes, albeit the power factor
decreases substantially. Clearly it would be advantageous if the apparent
motor current were to reduce in approximate relation with load decrease,
while at the same time the power factor remains about constant. Without
dynamic control of the motor operating conditions, such relatively
constant power factor operation is unattainable in induction motors of
ordinary (and economimcally cheap) design. My instant invention now
teaches a controller which can expedite such a desirable characteristic
from virtually any cheap motor design through the mere inclusion of two
sets of RUN windings, one of which is constantly excited to provide
sufficient magnetic field flux to drive the motor's rotor under reduced
load conditions, while the other RUN winding is increasingly excited as
the load increases. The combining of the separate magnetic fields produced
by the two RUN winding sets serves to provide a variable operating flux
density in the motor which is just sufficient to keep the motor running
without stalling under any load condition between that of reducd load and
full load, while at the same time economizing on the use of electrical
energy. The inclusion of the second set of RUN windings in even cheap
motors such as the aforementioned General Electric `Form V` or a `Form J`
style imposes no significant manufacturing difficulty because the meter
inclusion of a second set of RUN windings is little different from the
manufacturing practice involved in winding separate START and RUN windings
in the first place: e.g., the maker merely winds three sets of windings
(with the two RUN windings being wound with somewhat lighter gauge wire)
instead of the usual two winding sets.
DESCRIPTION OF DRAWINGS
My invention is illustrated with nine sheets of drawings showing twelve
figures.
FIG. 1 - Functional diagram showing principal elements which comprise my
invention.
FIG. 2 - Waveforms depicting signals which are essential for understanding
the advantageous performance of my invention.
FIG. 3 - Operational diagram showing general circuitry configuration which
enables practice of my invention.
FIG. 4 - Waveforms depicting signals which are found in the circuitry
configuration shown in FIG. 3.
FIG. 5 - Modification of general circuit configuration of FIG. 3 to include
a memory device which modifies the controller's dynamic characteristics.
FIG. 6 - Graphical plot of change in thyristor gate control signal delay
relative with motor current phase lag.
FIG. 7 - Hookup of electrical elements of a typical refrigerator with my
invention's control circuits.
FIG. 8 - Circuit for providing limited thyristor control range for the
secondary RUN winding.
FIG. 9 - Circuit which adapts the circuit of FIG. 8 to give a different
thyristor phase delay control range.
FIG. 10 - Waveforms depicting signals which are found in the circuitry
configurations of FIG. 8 and FIG. 9.
FIG. 11 - Circuit which adapts the circuit of FIG. 8 to skew the thyristor
phase delay control range even into the next half cycle.
FIG. 12 - Inductive pickup produces a sample of the RUN windinging lagging
current phase.
DESCRIPTION OF MY INVENTION
My invention involves the use of an induction motor of ordinary commercial
design which has been engineered to include two (or more) RUN winding
sets. Ordinarily, a main RUN winding set is coupled directly with the a.c.
power source, while a secondary RUN winding set is variably coupled
through a thyristor with the a.c. power source. The main RUN winding set
is predetermined to have sufficient ampere/turn capacity to produce the
flux density to excite the motor field and achieve normal operation for
light motor loads. The secondary RUN winding set is further predetermined
to have sufficient additional ampere/turn capacity to produce additional
magnetic field flux density which adds with the main RUN winding set flux
density so as to obtain reliable operation of the motor in the range
between that of a light load, which is excited by the main RUN winding set
alone, up to a condition of full motor load. The main RUN winding set
ordinarily is of more substantial construction, thereby providing a
greater portion of the motor's total magnetic field excitation. A.c. phase
control, or variable-phase power control as referred to in my invention is
used in the general context which is more particularly explained in
technical literature, such as in the General Electric Co. (Syracuse, N.Y.)
"SCR Manual", 4-th Edition, Edited by F.W. Gutzwiller (especially sections
9 and 10).
In FIG. 1 the a.c. induction motor 10 is provided with a rotor 10-1 (that
functionally drives a mechanical load which is not shown), a START winding
10-2, a main RUN winding 10-3, and a secondary RUN winding 10-4. In
addition, the motor may include a `start` capacitor 10-21 and a
centrifugal `start` switch 10-22. The main RUN winding couples directly
with the a.c. power line L1, and through the current phase sensor 40 with
power line L2. Thus, the main RUN winding is fully excited by the a.c.
line power.
A voltage phase sensor 30 couples via lines 32-1, 32-2 with the a.c. power
lines L1, L2 and produces a `voltage phase` signal on line 34 that couples
with the input PE of the phase detector 50. Current flowing through the
main RUN winding 10-3 also couples through the current phase sensor 40
(via lines 42-1, 42-2) which produces a lagging `current phase` signal on
line 44 that couples with the input PI of the phase detector 50.
The phase detector 50 determines the phase difference between the `voltage
phase` (reference) signal and the lagging `current phase` (error) signal,
producing a phase difference signal on line 52 that couples with a
proportional controller 60. The proportional controller functions to
determine a range of proportional power control signals on line 62 in
response to changes in the phase difference signal provided on line 52.
The proportional controller usually determines the outputted signal on
line 62 to have a larger dynamic range of electrical degrees of change
than what is presented on the input signal line 52. It is common that an
induction motor may have a range of lagging current which extends between
about -20 degrees and -60 degrees (for example). Meanwhile, it is usually
desirable that the thyristor 20 be enabled to phase control the power
coupled with the secondary RUN winding 10-4 over a much wider range: say
from about -0 to -180 degrees. In fact, in may practical motor
applications it may be desirable to obtain the full 180 degree control of
the thyristor gate delay with a mere 10 to 20 degree variation in motor
current phase leg. How this expanded phase control variation is produced
is one of the important aspects of my invention which shall be further
explained.
A d.c. power supply 70 provides a source of low d.c. voltage 72-1, 72-2 for
operation of the attendant electronic circuits which comprise the
operational circuits that make up my invention.
In FIG. 2 I depict some waveforms which may give better understanding of my
invention's central improvement. The waveform AE is typical of the kind of
a.c. power control afforded by the earlier teachings of the mentioned Nola
and Chang et al patents. You will see that the abrupt phase controlled
turn-ON portion AEAA of the waveform AEBA gives rise to sudden changes in
a.c. line conditions, and that these changes occur for every power
half-cycle such as shown AEAB for the other half-cycle AEBB. When the
phase-delay is even greater (in excess of -90 degrees), the half-cycle
waveforms AEDA, AEDB are shown to have even shorter and more severe
turn-ON AECA, AECB pulse transistions which cause the sudden power changes
reflected into the a.c. power line to look almost like `spikes`. The
result of such operation is power loss caused by the severe harmonic
distortion of the power line waveforms, resistance losses due to the high
current surge once turn-ON occurs, electrical noise, `flickering` of
lights connected with the same power line, and other undesirable effects.
With my invention, the main RUN winding (such as winding 10-3 of FIG. 1) is
fully excited by the a.c. waveform RWA of FIG. 2. This results in a
symmetrical power flow RWAA, RWAB having negligible distortion. The
controlled secondary RUN winding (such as winding 10-4 of FIG. 1) is
partially excited by a thyristor controlled power flow represented by
waveform RWB. In this case, the half-cycle waves are abruptly turned-ON
RWBA, RWBB by the thyristor, and when sufficiently delayed may even appear
like short `spikes` RWBC, RWBD. When combined with the steady, full-cycle
power draw RWA of the main winding, the controlled power flow RWB is
merely a portion of the total power flow, as shown by waveform RWC. The
composite waveform RWCA, RWCB represents that of the steady power draw RWA
combined with the controlled power draw RWB. In a like way, the waveform
portion RWCC, RWCD shows the combination of the steady power draw RWA with
the abrupt spikes RWBC, RWBD of waveform RWB. You should take particular
note of the minimal effect the phase controlled secondary RUN winding
power draw has on the overall waveform character, as shown by the
composite waveform RWC. The improvement is surprising and leads to a
remarkable increase in operating efficiency of my invention over that of
the prior art.
A circuit overview for my invention appears in FIG. 3 and includes an
induction motor 10 and the attendant RUN windings 10-3, 10-4. The main RUN
winding 10-3 couples through a resistor 10-31 to the a.c. power line: the
result is a voltage drop across the resistor having a phase relationship
which mirrors the lagging current flow through the main RUN winding. The
secondary RUN winding 10-4 couples with a triac 120, which includes a gate
122 that may turn the triac ON to obtain power flow from the a.c. power
lines L1, L2. The capacitor 126-1 and resistor 126-2 operate as a snubber
network to protect the triac against problems which may arise due to
fast-rising voltage changes which may occur when the driven load appears
inductive (e.g., fast dv/dt changes which can produce erratic
commutation). The additional set of switch contacts 10-41 operate in
concert with the `start` switch contacts 10-22 and therefore are closed
during motor starting. The contacts serve to bypass the heavy start-up
current rush around the triac 120. This novel arrangement protects the
triac from unecessary abuse and enables the use of a motor economical,
smaller rated triac because it has to merely handle the secondary RUN
winding current when the motor is properly running, and not the excess
current drawn during motor starting.
Inputs `A` and `B` of a voltage zero-cross detector 130 couple essentially
with the a.c. line voltage (waveform E of FIG. 4) which appears across the
line terminals L1, L2. A brief pulse signal (waveform XE of FIG. 4) is
produced on line 134 each time the voltage waveform EA, EB goes through
zero EXA, EXB (e.g., two pulses per cycle). The voltage pulse on line 134
couples with the SET input of a flip-flop latch 150, which when `set`
produces a HIGH logic level on the Q output
Inputs `A` and `B` of a current zero-cross detector 140 couple with the
a.c. voltage signal (waveform I of FIG. 4) which develops as a voltage
drop across resistor 10-31 due to (lagging) current flow in the run
winding 10-3. As a result, two pulses (waveform XI of FIG. 4) are produced
on line 144 for each a.c. current cycle. As depicted in the waveform XI,
the pulses XIAA, XIAB coincide in time with the less-lagging current
waveform IAA, IAB zero-cross coincidence IAXA, IAXB. When the current lags
more (as brought on by reduced power factor, or lighter motor loading) as
shown by waveform IBA, IBB the zero cross pulses XIBA, XIBB shift in
relative time to coincide with the zero-crossover coincidence IBAX, IBXB
of the current waveform. In practice of my invention, the timing of the
current zero-cross pulses XIBA, XIBB constantly shift about in time
relative with the voltage zero-cross pulses XEA, XEB. The pulses produced
on lien 144 then couple with the RESET input of the flip-flop 150, and
when reset has ocurred a LOW logic level appears on the Q output. As shown
by waveform LQ of FIG. 4, the latch 150 /Q output 152 signal (waveform LAA
of FIG. 4) is set LOW by the voltage zero-cross, is held low for a period
of time, and then returns HIGH (depicted by waveform LBA1, LBB1 or LBA2,
LBB2 of FIG. 4) when the current zero-cross pulse XIAA, XIAB (or XIBA,
XIBB) occurs.
The /Q output 152 from latch 150 couples with the RESET input of a delay
counter 160. The delay counter is clocked from pulses on line 182-1 and is
predetermined to count-up to a preestablished value on bus 160-1 which
produces a LOW output from the decoder 162 on line 164 that couples with
the RESET input of counter 166. It is my intent that counter 160 `delays`
the onset of counter 166 operation for a brief period of time which
coincides with the least value of lag delay (e.g., highest power factor)
which may be reasonably expected from the induction motor load. When RESET
of counter 166 goes `low` the counter will be clocked and advance by 0 to
15 counts (for example) depending upon the time which lapses between the
decoder producing a LOW signal on line 164, and the occurrance of the next
zero-cross current pulse. If current lag is relatively large (as with a
light motor load) counter 166 advances further than what has time to do if
the motor is more heavily loaded and the current lag is less. Waveform CP
of FIG. 4 shows the relationship between the clock pulses CPN and the
variations in timing of the current signal produced reset of latch 150 as
depicted by waveforms LBA1 and LBA2.
The advanced states on the counter 166 output bus 166-1 couple with the
DATA input of an edge triggered latch 170. The input data is thus
transferred to the output bus 170-1 the instant the /Q output of latch 150
goes HIGH. Latch 170 thusly serves to store the most recent count value
while the counter 166 is recounting during the next time period. In order
to produce symmetrical a.c. power control (which acts to reduce harmonic
distortion and line imbalance losses) I provide that each controlled a.c.
power cycle is self-completing: i.e., the first and second half-cycles are
of about the same duration. A divide-by-two flip-flop 158 is clocked from
the latch 150 Q output LOW-to-HIGH transistions, producing a pulse on line
158-1 the transistion of which is in-phase with the a.c. power voltage
phase. The resulting pulse serves the CLOCK the edge-triggered data latch
172 that effectively transfers the byte signal on bus 170-1 to bus 172-1
once during every a.c. power voltage phase cycle.
A clock 180 produces a relatively high-frequency signal which is
divided-down by the counter 182, producing a signal of necessary frequency
to clock counters 160 and 166. In addition, a lower-frequency signal (say
1,920 hertz for 60 hertz a.c. power) couples with the CLOCK input of a
4-bit counter 184. The RESET input of the counter couples with the voltage
zero-cross pulse signal on line 134, and is therefore reset at the onset
of each a.c. power half-cycle. After reset, the counter quickly advances
from count `0` to count `15` (binary 0000 to binary 1111) on bus 186. A
comparator 174 receives an A-IN signal from the latch 172 output data bus
172-1 and a B-IN signal from the counter 184 output bus 186. When the
counter 184 `counts-UP` to a binary weight signal that equates to the
stored binary signal appearing on bus 172-1, coincidence of A=B in the
comparator produces a HIGH pulse signal on output 174-1 that couples
through the triac driver 124 to produce a signal on line 122 that operates
the gate of the thyristor 120 to effectively turn the thyristor ON during
the rest of the instant a.c. power half-cycle. Waveform TG in FIG. 4
depicts the thyristor gate turn-ON signal pulse timing relative with the
plural clock pulses CPN. To interpret this waveform, it shall be seen that
the pulse TGPA occurs at the onset of the a.c. power 180-degree half-cycle
when the signal on bus 172-1 is binary 0000 because coincidence occurs in
the comparator immediately at the start of the half-cycle. Conversely, if
the signal on bus 172-1 has a binary weight of 1111 then counter 184 must
count-UP 16 steps and thus coincidence in the comparator is delayed in
time until near the end of the half-cycle, as depicted by pulse TGPB. Of
course different byte signal weights on data bus 172-1 will produce pulses
having timing intermediate of TGPA and TGPD (i.e., laying between about 0-
and 180-degrees of electrical phase delay prior to thyristor 120 turn-ON.
A computer program can serve to develop the best clock frequency choices,
as determined by specifying a range of minimum and maximum current phase
delay (e.g., power factor range). Furthermore, the values can be optimized
to restrict the dynamic range of thyristor control. For example, the
values can be determined such that the thyristor always operates over a
range of phase delays considerably less than the full 0- to 180-degree
maximum capability. The following program may be utilized for such
operative selections:
__________________________________________________________________________
10 'DETERMINATION FOR CLOCK FREQUENCY OF PHASE-LAG COUNTER @:CFPLC.BAS
V1.0
20 '(c) H. Weber -=- K1VTW -=- 4/11/88 -=- CP/M-80 -=- MBASIC-80 -=- DEC
VT-180
30 GOSUB 650:GOSUB 660
35 PRINT:GOSUB 645
40 PRINT "THIS PROGRAM WILL QUICKLY DETERMINE THE CLOCK FREQUENCY FOR
THE"
50 PRINT "PHASE-LAG ACCUMULATOR COUNTERS RELATIVE WITH DIFFERENT POWER
FACTORS."
55 GOSUB 645
60 GOSUB 670
70 INPUT "LOAD Motor Model No. or Description: ",MN$
80 IF LEN(MN$)>30 THEN 30
90 PRINT:INPUT "Enter MAXIMUM LOAD Current Lag in Degrees ",PFA
100
IF PFA>90 OR PFA<0 THEN 90
110
PRINT:INPUT "Enter MINIMUM LOAD Current Lag in Degrees ",PFB
120
IF PFB>90 OR PFB<0 THEN 110
130
IF PFA>PFB THEN 90
140
PRINT:INPUT "Accumulator Counter Division Factor `n` ",NF
150
IF NF<2 OR NF>256 OR NF>INT(NF) THEN 140
160
PRINT:INPUT "Enter Power Line Frequency (Hertz) ",LF
170
IF LF<25 OR LF>400 THEN 160
180
PRINT:INPUT "Enter MINIMUM Thyristor | | |
