 |
Description  |
|
|
The present invention relates to motor control systems and, more
particularly, to phase angle or vector control of a permanent magnet
motor.
BACKGROUND OF THE INVENTION
In electromechanical systems requiring fast response and four quadrant
operation with good performance near zero speed, an electrical machine
must essentially provide controlled torque over a wide range of operating
conditions. Historically, a separately excited direct current (DC) motor
has been the primary machine type employed in such situations. The
proportional relationship between motor armature current and motor torque
provides a direct means of achieving torque control. Use of high frequency
DC choppers with current feedback provides direct control of current and
overcomes the problem of speed dependence caused by the armature circuit
counter emf. Excellent torque control can be achieved until the counter
emf becomes comparable to the chopper input voltage. Field weakening is
also available to allow operation in the high speed, constant HP range.
Developments in the theory of controlling alternating current (AC) machines
coupled with technological developments in power switching systems and
control electronics now provide capability for achieving controlled torque
operation of AC machines. As in the DC machine, torque control is obtained
by controlling the motor current. In the AC machine, however, this control
must be in terms of both amplitude and phase, which has led to the generic
term "vector control". In addition, unlike the DC machine where
orientation of field flux and armature mmf is fixed by the commutator and
brushes, the AC machine requires external control of the field flux and
armature mmf spatial orientation. Without such control, phase angles
between the various fields in an AC machine vary with load (and during
transients) giving rise to complex interactions and oscillatory dynamic
response. Control systems for AC machines which directly control these
space angles have come to be called "field orientation" or "phase angle"
or "vector" controllers. The term "field orientation" generally means a
system which attempts to produce a 90.degree. space angle between
specifically chosen field components (currents and/or fluxes) so as to
closely emulate a DC machine.
High frequency chopping and current feedback are used to obtain current
control and overcome the speed dependent counter emf in field orientation
control of an AC machine. A pulse width modulated (PWM) inverter with
current loop control has been the controller of choice, although voltage
control is feasible and other types of inverters are often used.
Notwithstanding the improvement in operating characteristics of the AC
machine using such phase angle or field orientation controllers, the
desire to provide a machine having the advantageous characteristics of the
DC machine has led to development of a brushless DC machine and, more
particularly, to a permanent magnet DC machine, hereinafter referred to as
an electronically commutated motor or ECM. In the ECM, there are provided
multiple field windings which are energizable in a selectable sequence to
establish a rotating magnetic field. A rotor, constructed of permanent
magnets, has a substantially constant magnetic flux orientation which
interacts with the rotating magnetic flux field of the field windings to
effect rotation of the rotor. A more detailed description of an ECM may be
had by reference to U.S. Pat. No. 4,005,347 to Erdman issued Jan. 25, 1977
and assigned to the assignee of the present application.
Control systems developed for the ECM have generally been PWM inverter
systems using either square-wave current or voltage control to regulate
motor torque. These systems typically require rotor position sensors to
inform the control electronics of the instantaneous rotor position to
insure proper energization patterns for the stator windings to produce
rotation. Such rotor position sensors (e.g., encoders, resolvers, Hall
effect devices) are undesirable because of their cost, volume and
susceptibility to damage and failure. Alternatively, it is possible to
eliminate the need for these sensors by measuring the back emf voltages
generated by the spinning rotor magnets to determine rotor position. An
exemplary control system of this latter type is disclosed in U.S. Pat. No.
4,654,566 to Erdman, issued Mar. 31, 1987 and assigned to the assignee of
the present invention.
While the ECM using the aforementioned PWM control techniques has resulted
in an electromechanical system which combines many of the characteristic
advantages of the DC machine with the low maintenance and high speed
capability of the AC machine, it is desirable to achieve similar
performance characteristics in other types of permanent magnet machines
which are supplied with sinusoidal voltage or current excitation instead
of square-wave excitation.
FIG. 2 is a perspective view of one form of ECM or permanent magnet (PM)
motor M with multiple stator windings. As illustrated, the rotor is
removed from the stator. An illustrative three-phase winding for motor M
is shown in FIG. 3A. Rotor 15 is constructed with alternating magnetic
polarity magnets shown in what is sometimes referred to as a
surface-magnet construction. While this construction is very simple and
well known, it is also possible to construct such rotors in what is known
as "interior magnet" form to create an interior permanent magnet (IPM)
motor. As shown in FIG. 3, the magnets in the IPM motor are mounted below
the rotor surface and overlaid with a magnetic material which serves to
protect the magnets and strengthen the rotor. While this provides an
improved structure, it creates special control problems in that the
magnetic material is susceptible to induced magnetic flux fields from the
motor stator winding currents. Furthermore, the IPM motor requires
sinusoidal excitation for smooth torque production. Thus operation of this
type of sinusoidally-excited PM machine without a rotor position sensor
produces special control problems since the back-EMF cannot be directly
detected at the motor terminals as in a square-wave ECM drive due to the
inducted magnetic flux from the stator current causing an addition to the
back EMF.
The sinusoidally-excited permanent magnet motor also demonstrates a
start-up problem not experienced in field orientation control of AC
induction motors if there is no position sensor. In an AC induction motor,
the magnetic flux in the rotor can be initially established in an
arbitrary orientation by excitation of the stator windings with a current
of predetermined orientation. The rotor of the induction motor is
symmetrical about its axis so that there is no preferred rotor excitation
at startup. In the permanent magnet motor, the permanent magnets establish
a fixed orientation for the rotor field excitation flux. The applied
stator current must be properly oriented with respect to the rotor magnets
to generate torque. This poses a problem since orientation of the magnets
is not known at startup. Without some means of determining the rotor
orientation, precise torque control at start-up cannot be achieved in the
absence of a rotor position sensor.
SUMMARY OF THE INVENTION
It is a general object of the present invention to provide a control system
and method for a sinusoidally-excited PM motor which permits precise
control of machine torque.
It is a more specific object to provide a method and control system for
implementing field orientation or phase angle control of a
sinusoidally-excited PM motor without using a rotor position sensor.
It is another object to provide a field orientation control system for a
sinusoidally-excited PM motor which overcomes the special problems of
drive startup without using a rotor position sensor.
In one embodiment, there is provided a control system for a permanent
magnet motor having a stationary assembly, a plurality of winding stages
associated with the stationary assembly, and a permanent magnet rotor
arranged in selective magnetic coupling relation with the winding stages
for rotation in response to energization of the winding stages in the
preselected sequence. The control system comprises monitoring means
connected for sampling voltage and current supplied to the winding stages
and producing signals representative thereof, a three-phase to two-phase
converter means for converting the voltage and current signals to
corresponding two-phase rotating vector signals, power inverter control
means for selectively energizing the winding stages in a sinusoidal
fashion to effect rotation of the permanent magnet rotor means, a vector
rotator means responsive to DC signals representative of quadrature and
direct axis current command signals and to the two-phase rotating vector
components for producing two-phase (.alpha.- and .beta.-axis) control
signals which are varying in synchronism with the two-phase vector
signals, and function generating means responsive to a torque command
signal for producing the direct and quadrature axis DC current command
signals. The synchronously varying .dbd.- and .beta.-axis signals are
supplied to the power inverter control means for effecting sinusoidal
energization of the permanent magnet motor. The motor is preferably an
interior permanent magnet motor and the function generators convert the
torque command signal into direct and quadrature current commands
according to predetermined functions. The control system includes means
for initially establishing a preselected amplitude of .alpha.-axis current
in the motor at start-up prior to establishing .beta.-axis current such
that the rotor is caused to become initially aligned with the selected
.alpha.-axis.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram showing the major components of a control system
in combination with an electronically commutated motor;
FIG. 2 is an exploded, perspective view of the main elements of a surface
permanent magnet motor;
FIG. 3 is a cross section view showing laminations, with permanent magnets
in place, of an interior permanent magnet motor which is controllable by
the control system of the present invention;
FIG. 3A is a schematic diagram showing the winding stages and terminals of
the motor of FIGS. 2 or 3;
FIG. 4 is a vector diagram illustrating flux, current and voltage vectors
in an interior permanent magnet motor;
FIG. 5 illustrates in block diagram form an exemplary system for
controlling an IPM motor in accordance with the present invention;
FIG. 6 is a simplified block diagram of one circuit for generating
estimates of the flux vectors in an IPM motor;
FIG. 7 is a simplified block diagram of one circuit for converting current
command signals to rotating vector signals synchronized to a motor; and
FIG. 8 illustrates exemplary waveforms of the relationships between
commanded torque T* and commanded d- and q-axis stator currents (I.sub.d *
and I.sub.g *) executed by function generators 61 and 63 in the system of
FIG. 5.
DETAILED DESCRIPTION OF THE EMBODIMENT
FIG. 1 schematically illustrates a PWM motor control system which includes
a permanent magnet synchronous motor M adapted to be energized from a DC
power source and having (see FIG. 2) a stationary assembly including a
stator or core 13 and a rotatable assembly including a permanent magnet
rotor 15 and a shaft 17. The motor laminations, as shown in FIG. 3,
include conventional stator laminations 19 containing winding slots 21, as
typically employed, for example, in polyphase induction motors. Rotor
laminations 23 are one-piece laminations mounted on a shaft through
central opening 27. The rotor laminations contain slots 25 through which
permanent magnets are passed, when the slots are aligned, with north and
south poles of the magnets oriented as indicated by the N and S
designations, respectively, in the figure. Stator 13 includes a plurality
(e.g., three) of winding stages S1, S2 and S3 (FIG. 3A) adapted to be
electronically commutated in at least one preselected sequence, e.g.
A,B,C, although the invention is not limited to having three winding
stages.
When the winding stages S1, S2, S3 are sinusoidally-energized in an
appropriate temporal sequence, a radial magnetic flux is established which
moves clockwise or counterclockwise around the stator bore, depending on
the preselected sequence or order in which the stages are energized. This
moving field intersects the magnetic flux developed by the permanent
magnet rotor to cause rotor 15 to rotate relative to stator 13 in the
desired direction to develop a torque which is a direct function of the
intensities or strengths of the magnetic fluxes. This type of PM machine
is well known to those skilled in the art of motor drives.
Further, while PM synchronous motor M is illustrated herein for purposes of
disclosure, it is contemplated that other such motors of different
constructions and/or different winding arrangements may be utilized in one
or another form of the invention so as to meet at least some of the
objects thereof.
The winding stages of motor M are commutated energized in synchronism with
the rotor position by sensing the rotational position of the rotatable
assembly or rotor 15, as it rotates within the bore of stator 13, and
utilizing electrical voltage and current signals generated as a function
of the rotational position of the rotor to sinusoidally excite the winding
stages in different preselected orders or sequences that determine the
direction of the rotation of the rotor. If square-wave current pulses are
used to excite the motor stator windings instead of sinusoids, position
sensing may be accomplished by a position-detecting circuit responsive to
the back-emf of the ECM to provide a simulated signal indicative of the
rotational position of the rotor to control the timed sequential
application of voltage to the winding stages of the motor. Such a system
is disclosed in U.S. Pat. No. 4,654,566 to Erdman, issued Mar. 31, 1987
and assigned to the assignee of the present invention. However, this
scheme cannot be used if sinusoidal excitation of the motor is desired or
required, and inductance of the motor of FIG. 3 is often too high (due to
the extra iron above the magnets) so that the open phase voltage signal
required is not available.
Referring back to FIG. 1, power supplied from a 115 V 60 Hz AC line or
other suitable source is rectified by a rectifier circuit 29 which defines
a DC power source and is applied to a power inverter switching circuit 31
which constitutes means for controlling application of the DC voltage to
the winding stages to provide the winding stages with a resultant
effective sinusoidal voltage. Power switching circuit 31 responds to a set
of control signals from a control signal circuit 33 for exciting the
winding stages by pulse-width modulating the DC voltage to produce a
balanced three-phase sinusoidal excitation for the motor. The set of
control signals produced by control signal circuit 33 are a function of
rotor position which is derived from a position sensing circuit 35,
selected motor parameters, and the applied torque command.
The description thus far has been directed generally at control system
requirements for a permanent magnet synchronous motor. More specifically,
FIG. 4 illustrates a typical operating condition for an interior permanent
magnet motor, and is a vectorial representation of the motor magnetic flux
and the vectorially resolved motor current and voltage. The rotor is
simplified to a two-pole representation with the magnets aligned along the
direct or d-axis. The quadrature or q-axis is positioned to be along the
axis of the no-load counter EMF. The d and q axes are derivations of the
dq equivalent circuit model of a motor, an example of which is given in
U.S. Pat. No. 4,245,181 to Plunkett, issued Jan. 13, 1981 and assigned to
the assignee of the present invention. Under load, the voltage vector
V.sub.e shifts to position A. The current vector I.sub.e lags the voltage
vector V.sub.e for optimum efficiency and torque production (the vectors
are shown with respect to the rotating rotor 15 so that each is
essentially fixed with regard to the d-axis). The position and amplitude
of the voltage vector V.sub.e can be derived from the three applied stator
voltages. Since these voltages are chopped waveforms, the unfiltered
V.sub.e signal tends to be chopped and noisy as well. However, a measure
of the amplitude and position of the voltage vector can be developed by an
integration process which filters some of the chopping-generated ripple.
FIG. 5 is a block diagram of a control system, in accordance with the
present invention, for controlling permanent magnet motor M using field
orientation control techniques responsive to a torque command signal T*.
In this system, motor M is driven by a sinusoidal PWM inverter 37 which
applies current to the motor windings in at least one preselected sequence
to effect rotation of the motor rotor. A current feedback loop 39 provides
signals from motor M to a current control circuit 41. Circuit 41 provides
gating signals to power switching devices (not shown) of inverter 37 in a
manner to regulate current in motor M to match the commanded i.sub..alpha.
* and i.sub..sym. * current values. The .alpha.- and .beta.-axis
components represent two-phase equivalent values for the three-phase
stator A, B, C phase values. A more complete description of a PWM inverter
and current control circuit is given in Young U.S. Pat. No. 4,642,537,
issued to the instant assignee Feb. 10, 1987, the disclosure of which is
hereby incorporated by reference.
Current control circuit 41 is responsive to control signals i.sub..alpha. *
and i.sub..beta. * which are AC signals sinusoidally varying at the
frequency of rotation of the motor rotor and in synchronism therewith.
These control signals are effective to cause generation of the gating
signals on lines 43 in the appropriate sequence to energize the switches
within inverter 37. Signals I.sub..alpha. * and I.sub..beta. * are
generated by a vector rotator 45, which converts direct and quadrature
axis current commands I.sub.d * and I.sub.g * into rotating or AC
quantities in response to .vertline..psi..vertline. sin .omega..sub.e t
and .vertline..psi..vertline. cos .omega..sub.e t signals referred to
subsequently as .psi..sub..alpha. .omega..sub..beta., respectively. The
excitation frequency .omega..sub.e is related to the rotor frequency
(speed) .omega..sub.r according to .omega..sub.e =P/2 .omega..sub.r where
P is the number of motor poles. The .psi..sub..alpha. and .psi..sub..beta.
signals represent the stator flux vector signals obtained by integration
of motor terminal voltage and current. A stator flux vector integrator
circuit 47 generates the flux vector signal .psi..sub..alpha. and
.psi..sub..beta. from measured terminal voltage and current of the motor.
A more detailed description of vector rotators and flux integrators may be
had by reference to L. Garces U.S. patent application Ser. No. 839,203
filed Mar. 13, 1986, now U.S. Pat. No. 4,677,360 issued June 30, 1987 and
assigned to the present assignee. Another description may be obtained by
reference to U.S. Pat. No. 4,388,577 issued June 14, 1983.
One form of flux vector generator circuit 47 is shown in FIG. 6. The input
signals are measured values of terminal voltages V.sub.AC and V.sub.BC and
currents I.sub..alpha. and I.sub..beta. obtained in a manner well known in
the art. The voltages V.sub.AC and V.sub.BC are linearly transformed to
vector quantities V.sub..alpha. and V.sub..beta. by well-known means. The
circuit values Rs and Ls of blocks 49, 51,53 and 55 are representative of
measured values of stator resistance and inductance. The I.sub..alpha. and
I.sub..beta. signals are effected by these values (in the circuitry of
FIG. 6 as well as in motor M of FIG. 5) before being summed with the
V.sub..alpha. and V.sub..beta. signals. Integrators 57 and 59 integrate
the composite signals (V.sub..alpha. -I.sub..alpha. Rs) and
(V.sub..beta.-V.sub..beta. Rs) prior to further summation with signals
I.sub..alpha. Ls and I.sub..beta. Ls for producing .psi..sub..alpha. and
.psi..sub..beta., the estimated flux signals in the quadrature and direct
axes of the stator of motor M. The measured voltages and currents are sine
and cosine quantities as are the flux signals, whenever the rotor is
rotating.
The estimated flux signals are used in vector rotator 45 of FIG. 5, shown
in detail in FIG. 7. Here the direct and quadrature current command
signals (DC quantities) are multiplied by the rotating flux signals
.psi..sub..alpha. and .psi..sub..beta. (rotating quantities) to obtain the
I.sub..alpha. * and I.sub..beta. * command signals referred to with
respect to FIG. 5.
In operation, at no load the .psi..sub..alpha. and .psi..sub..beta. signals
are sine waves such that .psi..sub..alpha. is a maximum when the rotor
magnet is aligned with the stator phase A winding axis. As load is
applied, the .psi..sub..alpha. and .psi..sub..beta. signals shift in angle
similar to the total internal flux of the motor. When applied to the
vector rotator 45, shown in FIG. 5, these signals synchronize inverter 37
to motor M. The current command signals I.sub.d * and I.sub.g * are DC
signals which control amplitude of the component of motor current aligned
with the rotor magnet flux and the perpendicular quadrature component.
FIG. 8 illustrates typical examples of the relationship between the
commanded torque T.sub.e * and the current component commands I.sub.g *
and I.sub.d * executed by function generators 61 and 63, respectively.
These functional relationships are selected to optimize some aspect of
motor drive performance, such as enforcing unity power factor operation or
minimizing the required stator current for each torque command value
T.sub.e *. These functional relationships necessarily reflect the
parameters of the PM motor M. The particular functions illustrated in FIG.
8 are appropriate for an interior permanent magnet machine in which the
magnets are buried inside the iron rotor structure.
In order to initiate rotation of the system of the present invention, a dc
current is initially applied to the motor in the .alpha. axis, i.e., only
an I.sub..alpha. component of current is supplied. The magnitude of
current I.sub..alpha. is sufficient to cause the rotor to move into its
preferred alignment with the applied stator field. After the I.sub..alpha.
current is applied for sufficient time to permit rotor alignment, inverter
37 is energized to command current to ramp up in the .beta. axis which
generates a rotating MMF excitation wave in the airgap, producing torque
and initiating rotor rotation. As soon as the motor begins rotating, the
magnets begin generating back-emf voltages which can be sensed as terminal
voltages, allowing the flux calculator circuit 47 in FIG. 5 to begin
normal running operation. This method allows the motor to be self-aligning
without prior knowledge of rotor position.
What has been described is a method and system for controlling a permanent
magnet motor and, in particular, an interior permanent magnet motor using
a field orientation control system, without a position sensor, which
automatically synchronizes operation of an inverter to the motor rotor
position for start-up and thereafter compensates for non-linear response
of the motor to changes in torque commands. While the system has been
described in what is presently considered to be a preferred embodiment,
other variations and modifications will become apparent to those skilled
in the art. Accordingly, it is intended that the invention be interpreted
within the full spirit and scope of the appended claims.
* * * * *
|
|
 |
|
 |
Description |
|
