 |
Description  |
|
|
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a structure of a stepping motor and a
method of driving the stepping motor, and particularly, relates to a field
magnet used in the motor of this type.
2. Description of the Prior Art
Conventional stepping motors of the hybrid type are superior in thrust to
the motors of the other types under equal conditions in electric source
capacity and motor rating. However, those hybrid stepping motors have a
disadvantage in that it is difficult to perform high-speed driving because
of the influence of inductance. The disadvantage is described in detail
with reference to FIG. 2 which shows a first conventional technique to be
improved.
Consideration will be made about the inductance of hybrid stepping motors.
In regard to windings 3 and 4, the magnetic circuit of FIG. 2 has a
magnetic path in the direction of the arrow. The magnetic reluctance of
the circuit is determined mainly by the shape of gaps. The magnetic
reluctance R.sub.c as to the windings 3 and 4 is represented by the
equation: R.sub.c =R.sub.g1 +R.sub.g2, in which R.sub.g1 and R.sub.g2
respectively represent the magnetic reluctance in the gaps of poles A and
A of a stator 1. The inductance of the windings is in proportion to the
square of the number of turns N and in reverse proportion to the magnetic
reluctance, as represented by the equation:
##EQU1##
in which K is a constant.
For driving the hybrid motor at a high speed, it is necessary to make a
current for the windings rise rapidly so as to generate large thrust in a
short time. Accordingly, the electrical time constant represented by the
ratio of inductance to resistance of the windings has a serious meaning.
The electrical time constant of the conventional hybrid stepping motor is
of the order of several mS, and various methods have been heretofore used
for driving the motor at a high speed with respect to the electrical time
constant. In the following, the conventional methods are described in
brief.
One of the methods is such that a resistor is connected in series at the
outside to increase the value of resistance to thereby minimize the time
constant. However, the method has a problem in that the increase of the
consumption of electric power by the resistor causes troubles, such as the
lowering of efficiency in the stepping motor, the overheat of the motor,
and the like.
Another one of the methods is such that the number of turns of the windings
is decreased or the gaps are enlarged to increase the magnetic reluctance
to thereby reduce the inductance. However, the method has a problem in
that the thrust constant of the motor represented by a ratio f/I of the
thrust f (newton) to the motor current I (Ampere) is adversely affected
and becomes lower under the condition of the same electric source
capacity.
As a means for increasing the magnetic reluctance without deterioration of
thrust, a method in which a magnet is provided in the magnetic circuit as
to the windings, that is, in which a magnet is provided in the magnetic
path of the arrow of FIG. 2, is considered. However, in the case where the
magnet is merely provided in the path of the arrow, the change of flux
interlinkage with the windings according to the change of magnetic
reluctance of the gaps is reduced to lower the thrust constant, though the
magnetic reluctance is increased corresponding to the magnetic reluctance
of the magnet. Accordingly, the reduction in thrust can not be avoided
even in this method.
To solve the aforementioned problems, a motor having the structure of FIG.
3 has been considered. The structure is such that in order to reduce the
inductance, permanent magnets 13 and 14 to be provided in the series
magnetic circuit of the windings are mounted on the surfaces of poles, and
in order to enlarge the change of flux interlinkage with the windings 3,
4, 5 and 6, the polarity of poles of the magnets is inverted corresponding
to the pitch of the teeth of the poles, so that the reduction of thrust is
prevented. In the structure, however, a magnetic leakage path is formed
between adjacent poles of the permanent magnets to reduce effective flux
in the gaps, so that the flux interlinkage with the windings is reduced.
Accordingly, sufficient thrust cannot be attained even by this structure.
In order to reduce the flux leakage, the permanent magnets must be thinned
sufficiently relative to the pitch of the poles of the permanent magnets.
However, the magnetic reluctance of the magnets decreases as the thickness
thereof decreases. Accordingly, a problem arises in that the inductance of
the windings increases. Further, in the case where the pitch of the poles
of the permanent magnets are enlarged, the pitch of the teeth must be
enlarged with the enlargement of the pitch of the poles of the permanent
magnets. Accordingly, a problem arises in that the thrust constant is
lowered because the change of flux relative to the positional change of
the movable member is reduced.
Consequently, even in the case where the polarity of poles of the magnets
are inverted corresponding to the pitch of the teeth in order to enlarge
the change of flux, it has been difficult to make the reduction of
inductance compatible with the improvement of thrust constant.
An example of the conventional stepping motor in which resistance is
increased to thereby improve response is disclosed in Takashi Kenjo and et
al., "Principle and Application of Stepping Motor", Sogo Electronics
Publishing Co., Ltd., pages 180 to 182, February 1979, an example of the
conventional stepping motor in which a magnet is provided in the magnetic
circuit of the windings is disclosed in Japanese Patent Unexamined
Publication JP-A-60-200757, and an example of the conventional stepping
motor in which the polarity of poles of a permanent magnet is inverted at
a pitch equal to the pitch of teeth is disclosed in Osahiko Nagasaka,
"Study of Prototype PM Linear Pulse Motor", Magnetics Research of the
Institute of Electrical Engineers of Japan, MAG-85-130, 1985.
Consequently, the aforementioned first conventional technique has the
problem that the thrust constant is lowered with the attempt to reduce the
electrical time constant, because the technique is not under sufficient
consideration as to the relation between the thrust constant and the
electrical time constant in the hybrid stepping actuator.
In the following, a motor driving method according to a second conventional
technique will be described with reference to FIGS. 10A and 10B. In the
drawings, the reference numeral 101 designates a section of a motor, and
the reference numeral 102 designates a circuit for driving the motor. The
motor has a pair of stators 103, and a movable member 108. The stators 103
are provided with A-phase and B-phase windings 104 and 106, respectively.
The movable member 108 is provided with permanent magnets 109 disposed at
the opposite sides thereof. Each of the permanent magnets 109 has a
plurality of poles disposed at equal intervals at the pitch equal to the
pitch .lambda. of the teeth of the stators 103 so that the poles N and S
of each of the permanent magnets 109 alternate. The permanent magnets 109
are disposed in a manner so that one permanent magnet 109 being in
opposition to the A-phase stator is shifted in phase by an electric angle
of 90 degrees (1/4 of the teeth pitch .lambda.) relative to the other
permanent magnet 109 being in opposition to the B-phase stator. The drive
circuit 102 is provided with two groups of transistors 110, 111, 120 and
121, and 112, 113, 122 and 123. The one group of transistors 110, 111, 120
and 121 are connected to each other in the form of H through the A-phase
winding 104, while the other group of transistors 112, 113, 122 and 123
are connected to each other in the form of H through the B-phase winding
106, the one and the other transistor group being connected across a DC
source 131, as shown in FIG. 10B. The transistors are ON-OFF controlled by
a control circuit 130 which is supplied with a signal from a sensor
circuit 132 for sensing the position of the movable member 108.
In the motor, the direction of flux interlinked with the windings 104 and
106 is inverted in accordance with the position of the movable member 108
so that the direction of a current flowing in the windings 104 and 106 is
inverted corresponding to the change of flux to thereby generate thrust in
the movable member. The changes of flux interlinkage with the respective
windings 104 and 106 are shifted in phase by 90 degrees from each other
corresponding to the shifting of the position of the permanent magnets 108
and 109. Accordingly, if the currents of the A-phase windings 104 and
B-phase windings 106 are inverted in accordance with the respective phases
of the currents, unidirectional thrust can be generated in the movable
member continuously at any position thereof. The groups of H-connected
transistors in the driving circuit 102 are driven to operate in a manner
as follows. That is, in the one group of H-connected transistors
associated with the A-phase winding 104, the transistors 110 and 121 are
turn on at a certain point of time to pass a current from the terminal a
to the terminal b, whereafter the transistors 111 and 120 are turn on at
the next point of time to pass a current from the terminal b to the
terminal a reversely. Also the other group of H-connected transistors
associated with the B-phase winding 106 are driven to operate in the same
manner as the above-mentioned one group of transistors. The aforementioned
reversible operation of current flowing in the windings 104 and 106 can be
accomplished by the foregoing ON-OFF operation of the transistors. The
transistors are controlled by control signals from the control circuit
130. The control circuit 130 receives the position signal from the sensor
circuit 132 for sensing the position of the movable member 108 so as to
judge whether the current is to be inverted or not to thereby control the
current. Alternatively, the control circuit 130 may judge the inversion of
the current by itself to thereby control the current by so-called
open-loop control without using the position signal.
The motor of this type has an advantage in that only thrust can be enlarged
by minimizing the pitch of the teeth with the motor rating kept fixed,
because the magnitude of thrust is in proportion to the change of flux
interlinkage with the windings. Accordingly, a load can be moved at a high
speed by use of such a motor driving method. Accordingly, the motor of
this type can be suitably used as an actuator for feeding a head in a disk
drive or the like.
An example of the motor of this type is disclosed in Japanese Patent
Unexamined Publication JP-A-56-74080.
The aforementioned second conventional technique has an advantage in that
large thrust driving in a motor can be attained, but has a problem in that
the current flowing in the windings is reduced with the increase of
reactance and with the rising of the induced voltage when the motor
becomes into a high-speed running condition, because the technique is not
under sufficient consideration as to the maintenance of the large thrust
in the high-speed condition.
In the following, a field magnet used in stepping motors and linear pulse
motors according to a third conventional technique, will be described.
Generally, a linear pulse motor using a field magnet is formed as described
in Japanese Patent Unexamined Publication JP-A-56-74080. According to the
above Japanese Patent Unexamined Publication, a stator is constituted by
an elongated base and a plurality of permanent magnets fixed to the upper
surface of the base. The permanent magnets are magnetized to provide N and
S poles alternately in a direction of movement of an armature along the
stator.
As described above, since the poles of the field magnet in the conventional
motor are formed by magnetization at a fine pitch, so that large
magnetomotive force cannot be generated.
Accordingly, the thrust of the conventional motor is too small to deal with
a large load. This causes limitation in the purposes of use of the motors
or actuators of this type.
SUMMARY OF THE INVENTION
An object of a first aspect of the present invention to solve the problem
in the first conventional technique is to provide a hybrid stepping motor
in which the electrical time constant is reduced without lowering the
thrust constant by optimizing the position and form of permanent magnets
used in the motor.
The foregoing object of the invention is attained by a structure of the
stepping motor in which each permanent magnet is provided to face the gap
of a stator or a movable member, the polarity of poles of the permanent
magnet is inverted at a pitch equal to that of the teeth of the movable
member or the stator facing the permanent magnet, and slots are provided
in the permanent magnet at positions where the polarity of poles of the
permanent magnet is changed over.
In the aforementioned structure in which the polarity of poles of the
permanent magnet is inverted at a pitch equal to that of the teeth of the
movable member or the stator facing the permanent magnet, the following
operations are attained. In a first condition in which the phase of the
permanent magnet is made coincident with that of the teeth, the flux of
the poles of the permanent magnet at the short gap length side (at the low
gap magnetic reluctance side) is interlinked with the windings. In a
second condition in which the phase of the permanent magnet is shifted by
half a pitch relative to that of the teeth, the gap magnetic reluctance is
equal to the poles different in polarity of the permanent magnet so that
the flux is terminated between the poles different in polarity of the
permanent magnet and is not interlinked with the windings. In a third
condition in which the gap length for the poles of the permanent magnet is
reversed to the first condition, the flux in the reverse direction to the
first condition is interlinked with the windings. As described above, the
flux interlinkage with the windings which is zero in the first condition
widely changes corresponding to the relative positional change of the
permanent magnet on the basis of the condition where the flux interlinkage
with the windings is zero. For example, the condition of flux interlinkage
with the windings changes from the first condition to the first condition
again through the second condition, the third condition, and the second
condition again. That is, the flux interlinkage with the windings changes
widely between positive and negative on the basis of zero, successively.
Accordingly, the thrust of the motor becomes very large.
Further, the inductance of the windings is reduced by arranging the
permanent magnet in series to the magnetic circuit of the windings so that
the electrical time constant of the motor can be reduced.
Further, consideration should be made on the point that slots are provided
at the positions where the polarity of poles of the permanent magnet is
changed over. That is, the slots are provided so that the plurality poles
of the permanent magnet are shaped in the form of teeth to thereby reduce
flux leakage between adjacent poles different in polarity of the permanent
magnet so as to increase flux in the gap. If such slots are not provided,
magnetic reluctance at the polarity changeover positions is so small that
flux concentrates at those positions so as to greatly increase flux
leakage to thereby reduce flux interlinkage with the windings. Otherwise,
if those slots are provided, the flux of the permanent magnet itself
decreases with the decrease of the area of the permanent magnet but the
magnetic reluctance between adjacent poles of the permanent magnet
increases due to the slots between adjacent poles different in polarity,
so that magnetic leakage is greatly reduced. Consequently, the provision
of those slots brings the increase of flux interlinkage with the windings
and the increase of thrust. Instead of provision of those slots, a
plurality of magnets may be provided at equal intervals so that poles
different in polarity are arranged alternately. Also in this case, the
same effect can be attained. In addition, the gaps or slots may be filled
with a non-magnetic material.
In the aforementioned second conventional technique, a current I which can
be made to flow in the one-phase windings of the actuator with an electric
source voltage E.sub.s is expressed by the equation:
##EQU2##
where r represents the resistance, L represents the inductance, v
represents the velocity, p represents the pitch, and k.sub.e represents
the induced voltage constant. In the equation ( 1), the reactance X is
expressed by the following equation:
##EQU3##
Accordingly, the thrust f generated in the one phase is expressed by the
equation:
##EQU4##
where k.sub.f represents the thrust constant.
On the basis of the equations, the current in the windings decreases and
the generated thrust decreases as the speed increases.
FIG. 11 shows a velocity-thrust graph of the motor. In the drawing the
broken line shows characteristics due to a term in the numerator of the
equation (2). The thrust linearly decreases as the induced voltage k.sub.e
.multidot.v increases. Further, the solid line shows characteristics due
to the term of reactance 2.pi.(v/p) L in the denominators of the equation
(2). The degree of decrease of the thrust is more intensive in the
low-speed area as shown by the solid line.
Accordingly, in spite of the fact the motor has a large thrust constant,
the current in the windings is reduced in the high-speed operation so that
sufficient acceleration cannot be attained.
Further, as described above, the conventional motor having the structure
that poles are shaped in the form of teeth has the advantage in that the
thrust (thrust constant) can be enlarged by minimizing the pitch of the
teeth. However, as the pitch of the teeth decreases, angular frequency
2.pi.(v/p) of reactance increases in spite of the same speed. Accordingly,
the effect of large thrust cannot be expected in the high-speed operation.
An object of a second aspect of the present invention to solve the problem
in the second conventional technique is to provide a method of driving a
motor which can prevent the lowering of the generated thrust with the
lowering of the current flowing in the windings in the aforementioned
high-speed operation.
The foregoing object of the second aspect of the invention is attained by
the driving method in which a control means is provided for dividing the
windings for each phase of the motor into two or more sections to thereby
change the number of turns of windings into which a current is made to
flow, and in which the speed of the motor is detected so that the number
of turns of the windings into which a current is made to flow is changed
in accordance with the motor speed.
In the following, the operation based on the aforementioned arrangement is
described.
In the motor, the windings for each phase is divided into two or more
sections, and a drive circuit is provided to pass currents through the
divisional winding sections independently. Further, means for detecting
the speed of the motor is provided so that the number of turns of the
windings through which a current is made to flow can be changed by the
control circuit in accordance with the speed of the motor. Accordingly,
the number turns of the windings into which a current is made to flow is
increased in the low-speed operation, so that high acceleration can be
attained by the increase of the thrust constant, while the number turns of
the windings into which a current is made to flow is reduced in the
high-speed operation, so that the decrease of current can be prevented to
maintain necessary thrust. Thus, it is possible to realize an actuator
which can maintain necessary thrust in a necessary range of speed and can
operate at a high speed.
An object of a third aspect of the present invention to solve the problem
in the third conventional technique is to provide a field magnet which is
suitably applied to a stepping motor so that large thrust can be attained.
Another object of the third aspect of the invention is to provide a field
magnet which is so simple in construction that workability in
manufacturing can be improved greatly.
The foregoing objects of the third aspect of the invention is attained by a
field magnet comprising a magnet substrate having magnet holding through
holes or slots disposed at equal intervals of a predetermined pitch and
magnetized in the direction of thickness, and permanent magnets magnetized
in the reverse direction to the magnet substrate and inserted into the
magnet holding through holes or slots of the magnet substrate to form the
field magnet in combination.
According to the aforementioned construction, a magnet substrate and
permanent magnets respectively magnetized in advance are combined with
each other. Accordingly, there is no demagnetization due to magnetization,
so that the magnetomotive force of the field magnet can be improved.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a sectional view showing an embodiment according to the first
aspect of the present invention;
FIGS. 2 and 3 are sectional views of the structure of a first example of
the conventional motor;
FIG. 4 is a view of an equivalent magnetic circuit of the embodiment
according to the first aspect of the invention;
FIG. 5 is a sectional view of another embodiment according to the first
aspect of the invention;
FIG. 6 is a front view of a rotary stepping motor;
FIG. 7 is a sectional side view of the rotary stepping motor;
FIG. 8 is a partly enlarged view of the shape of a teeth portion showing a
further embodiment according to the first aspect of the invention;
FIG. 9A is a sectional view of an actuator as an embodiment according to
the second aspect of the invention;
FIG. 9B is a circuit diagram of the actuator;
FIG. 10A is a sectional view of a second example of the conventional motor;
FIG. 10B is a circuit diagram of the second example of the conventional
motor;
FIG. 11 is a velocity-thrust characteristic graph of the conventional
motor;
FIG. 12 is a velocity-thrust characteristic graph of the embodiment
according to the second aspect of the invention;
FIG. 13 is a velocity rising characteristic graph in the second aspect of
the invention;
FIG. 14 is a perspective view of a field magnet showing an embodiment
according to the third aspect of the invention;
FIG. 15 is a perspective view of a conventional field magnet;
FIG. 16 is a principle view of a linear motor to which the third aspect of
the invention is applied;
FIGS. 17 and 18 are views showing a method for magnetizing the conventional
field magnet;
FIG. 19 is a development view showing the process of assembling the field
magnet depicted in FIG. 14;
FIGS. 20 and 21 are perspective views showing other embodiments according
to the third aspect of the invention; and
FIG. 22 is a sectional view showing an assembling machine for forming the
field magnet depicted in FIGS. 14 and 21.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
An embodiment according to the first aspect of the invention will be now
described with reference to FIG. 1.
Stators 1 and 2 are provided to face each other with a gap therebetween. A
movable member 7 is provided within the gap between the stators 1 and 2
and in parallel to the stators 1 and 2 to form predetermined gaps with
respect to the stators 1 and 2. Permanent magnets 8 and 9 are mounted on
the respective surfaces of the movable member 7 facing the stators 1 and 2
respectively. Each of the permanent magnets 8 and 9 has a plurality of
poles disposed at equal intervals and providing S and N polarities
alternately such that a pair of S and N poles appear repeatedly at every
distance corresponding to one pitch of the teeth of the stators 1 and 2.
The phase of the polarity of the permanent magnet 9 is shifted by
90.degree. (1/4 pitch of the poles of the same polarity of the magnet)
relative to the phase of the polarity of the permanent magnet 8. The
stator 1 has two poles A and A and the stator 2 has two poles B and B. The
relation in phase of the polarities of those poles of the stators are set
to be as follows. That is, when the one pole A approaches an N pole of the
permanent magnet 8, the other pole A is located near an S pole of the
permanent magnet 8 (or in other words, the phase of the pole A is shifted
by 2/4 pitch with respect to the phase of the pole A where one pitch is
corresponding to a distance between the leading edge of one pole of the
permanent magnet and the leading edge of the next pole of the same
polarity). Further, in this case, the pole B is located in the middle
between an S pole and an adjacent N pole of the permanent magnet 9 (or in
other words, the phase of the pole B is shifted by 3/4 pitch), and the
pole B is located in the middle between an N pole and an adjacent S poles
of the permanent magnet 9 (or in other words, the phase of the pole B is
shifted by 1/4 pitch).
A winding 3 is provided on the pole A of the stator 1, and a winding 4 is
provided on the reverse pole A of the same stator 1. A winding 5 is
provided on the pole B of the stator 2, and a winding 6 is provided on the
reverse pole B of the same stator 2.
The magnetic circuit of the aforementioned motor is shown in FIG. 4. In
FIG. 4, U.sub.11 represents magnetomotive force produced at the N-pole
side of the permanent magnet 8 with respect to the pole A, U.sub.12
represents magnetomotive force produced at the S-pole side of the same
permanent magnet 8 with respect to the pole A, U.sub.21 represents
magnetomotive force produced at the N-pole side of the same permanent
magnet 8 with respect to the pole A, and U.sub.22 represents magnetomotive
force produced at the S-pole side of the same permanent magnet 8 with
respect to the pole A. Similarly to this, U.sub.31 represents
magnetomotive force produced at the N-pole side of the permanent magnet 9
with respect to the pole B, U.sub.31 represents magnetomotive force
produced at the S-pole side of the same permanent magnet 9 with respect to
the pole B, U.sub.41 represents magnetomotive force produced at the N-pole
side of the same permanent magnet 9 with respect to the pole B, and
U.sub.42 represents magnetomotive force produced at the S-pole side of the
same permanent magnet 9 with respect to the pole B. R.sub.m represents
magnetic reluctance in each magnet, R.sub.g11 represents magnetic
reluctance in the gap of the pole A which receives flux from the N-pole
side of the permanent magnet 8, R.sub.g12 represents magnetic reluctance
in the gap of the pole A which receives flux from the S-pole side of the
permanent magnet 8, R.sub.g21 represents magnetic reluctance in the gap of
the pole A which receives flux from the N-pole side of the permanent
magnet 8, R.sub.g22 represents magnetic reluctance in the gap of the pole
A which receives flux from the S-pole side of the permanent magnet 8,
R.sub.g31 represents magnetic reluctance in the gap of the pole B which
receives flux from the N-pole side of the permanent magnet 9, R.sub.g32
represents magnetic reluctance in the gap of the pole B which receives
flux from the S-pole side of the permanent magnet 9, R.sub.g41 represents
magnetic reluctance in the gap of the pole B which receives flux from the
N-pole side of the permanent magnet 9, and R.sub.g42 represents magnetic
reluctance in the gap of the pole B which receives flux from the S-pole
side of the permanent magnet 9.
The magnetic reluctance in each gap changes sinusoidally with the pitch
.tau. of the teeth as one period according to the position of the teeth of
the movable member 7 relative to the teeth of the stators 1 and 2. When
the magnetic reluctance R.sub.g11 and R.sub.g22 are minimized, the
magnetic reluctance R.sub.g12 and R.sub.g21 are maximized. Accordingly, in
this case, the flux .phi..sub.11 and the flux .phi..sub.22 due to the
magnetic reluctance U.sub.11 and U.sub.22 are maximized, and the flux
.phi..sub.12 and the flux .phi..sub.21 due to the magnetic reluctance
U.sub.12 and U.sub.21 are minimized. Accordingly, the flux interlinkage
.phi..sub.1 with the windings 3 and 4 passes in the direction of the arrow
of FIG. 4. Contrariwise, in the case where the magnetic reluctance
R.sub.g11 and R.sub.g22 are maximized, the magnetic reluctance R.sub.g12
and R.sub.g21 are minimized. Consequently, in this case, the flux
interlinkage .phi..sub.1 with the windings 3 and 4 passes in the reverse
direction to the arrow of FIG. 4. As described above, the changes of flux
arise in the windings 3 and 4 according to the position of the movable
member 7. If a current flows in the windings, thrust is generated to move
the movable member linearly. On the other hand, the changes of magnetic
reluctance in the gaps at the lower side of the magnetic circuit of FIG. 4
are relatively shifted by the phase of .pi./2 compared with the upper side
of the magnetic circuit, so that thrust shifted by .pi./2 is generated.
Since the teeth pitch is selected to be a value of the order of
millimeters, the change of flux according to the positional change of the
movable member is so large that large thrust can be attained.
The magnetic reluctance R.sub.c of the magnetic circuit with respect to the
windings 3 and 4 is increased by the magnetic reluctance of the magnet and
is expressed by the following equation:
R.sub.c =(R.sub.g11 +R.sub.m)//(R.sub.g12 +R.sub.m)+(R.sub.g21
+R.sub.m)//(R.sub.g22 +R.sub.m)
Accordingly, the inductance of the windings 3 and 4 becomes smaller because
of the increase of the magnetic reluctance, and the inductance of the
windings 5 and 6 becomes also smaller, compared with that of prior art
windings.
Further, in this embodiment, slots are provided between adjacent poles
different in polarity in each of the permanent magnets to prevent flux
leakage from occurring between adjacent poles different in polarity. In
other words, each of the permanent magnets has teeth formed respectively
at poles, so that a gap is put between adjacent poles different in
polarity to thereby elongate the magnetic path to increase the magnetic
reluctance thereat. Accordingly, comparing with the magnetic reluctance of
the gap between the teeth which face the teeth of the permanent magnet and
which contribute to the change of flux interlinkage with the windings, the
magnetic reluctance between adjacent poles different in polarity of the
permanent magnet increases to thereby reduce the flux leakage between the
adjacent poles different in polarity. Accordingly, the permanent magnet
having such slots provided between adjacent poles does not reduce the
thrust constant, and therefore the permanent magnet can be thickened to
reduce the inductance suitably and the inductance can be reduced without
reduction of the thrust constant.
Further, the width of each slot of the permanent magnet may be larger than
that of the gap between the teeth faced by the slot. In this case, the
magnetic reluctance in the leakage path between adjacent poles of the
permanent magnet is larger than that of the gap, so that the
aforementioned effect can be attained more remarkably.
As described above, this embodiment can provide a linear stepping motor
having a large thrust constant and being superior in high speed
performance.
A second embodiment according to the first aspect of the invention is shown
in FIG. 5. In this embodiment, the aforementioned permanent magnets are
provided on the stators 1 and 2. Also in this embodiment, the same effect
as described above in the first embodiment, that is, the effect of large
thrust and high-speed driving performance can be attained. In addition,
this embodiment has another effect in that the permanent magnet can be
relatively reduced in size as well as weight compared with the first
embodiment, because the length of the permanent magnet is equal to that of
the pole of the stator, as measured in a direction of movement of the
movable member.
Although this embodiment has shown the case where the invention is applied
to a linear motor, it is a matter of course that the invention is
applicable to a rotary motor and that a high-speed and large-torque motor
can be provided by forming the rotary motor in the same manner as
described above.
FIG. 6 shows a further embodiment in which the invention is applied to a
rotary motor. FIG. 7 is a sectional view taken along the line VII--VII of
FIG. 6. In this embodiment, a permanent magnet 21 having poles which are
reversed in polarity at equal intervals in proportion of 1/2 of the pitch
of the teeth of the stator 19 in the same manner as the linear motor, is
mounted onto the outside of a rotor 20. Further, slots are provided at
positions where the polarity of poles of the permanent magnet 21 is
changed over. As shown in FIG. 7, the phase of the permanent magnet 21 in
the A-phase of the rotor 20 is shifted by 1/4 of the pitch .tau. compared
with the phase of the permanent magnet 25 in the B-phase thereof.
According to this structure, a large-thrust and low-inductance motor can
be attained in the same manner as the aforementioned linear stepping
motor.
FIG. 8 is a sectional view showing the structure of teeth in a further
embodiment. In this embodiment, a permanent magnet 28 is stuck to the
teeth portion 27 of a movable member or a stator by means of evaporating
deposition, sputtering or the like. In this embodiment, a fine teeth
structure can be formed corresponding to the shape of the teeth portion 27
by means of evaporating deposition or sputtering. Accordingly, this
embodiment has the effect in that the change of flux in accordance with
the positional change of the movable member becomes very large to thereby
increase the thrust constant.
In the following, the second aspect of the invention is described with
reference to FIGS. 9A and 9B.
In the drawings, the reference numeral 101 designates a section of a motor,
and the reference numeral 102 designates a motor drive circuit. The motor
101 has a pair of stators, and a movable member 108. The stators having
teeth facing a movable member 108 are provided with A-phase windings 134
and 135 and B-phase windings 136 and 137, respectively. The movable member
108 is provided with permanent magnets 109 disposed at the opposite sides
of the movable member 108 facing the stators. Each of the permanent
magnets 109 has a plurality of poles disposed at equal intervals at the
pitch equal to the pitch .lambda. of the teeth of the stators. The
magnetic position of one permanent magnet 109 on the A-phase side is
shifted by an electric angle of 90 degrees (1/4 of .lambda.) relative to
that of the other permanent magnet 109 on the B-phase side.
In the drive circuit 102, transistors 110, 111, 120 and 121 are connected
in the form of H through the A-phase winding 134. Transistors 111, 114,
121 and 124 are connected in the form of H through an A-phase winding 135.
Transistors 112, 113, 122 and 123 are connected in the form of H through a
B-phase winding 136. Transistors 113, 115, 123 and 125 are connected in
the form of H through a B-phase winding 137. Signals from a control
circuit 130 are applied to the bases of the respective transistors and a
signal from a position sensor 132 is applied to the control circuit 130.
In the following, the method of driving the motor is described.
Let each of the A-phase windings 134 and 135 of the motor 101 has the
number of turns which is half (N/2) the number of turns of the A-phase
winding 105 of the conventional motor. Similarly, let each of the B-phase
windings 136 and 137 has the number of turns which is half (N/2) the
number of turns of the B-phase winding 106. In the condition in which the
windings 134 and 135 and the windings 136 and 137 are respectively
connected in series, the rating and thrust of the motor are equivalent to
those of the conventional motor. Accordingly, the thrust constant k.sub.f,
the induced voltage constant k.sub.e and the inductance L are represented
by the following equations (3):
##EQU5##
In the case where the motor 1 operates at a low speed, the combination of
transistors 110 and 124 and the combination of transistors 114 and 120 in
the control circuit 102 are alternately turned on and off to pass a
current through the windings 134 and 135 through the points a, b, c and d
in order successively. On the B-phase side, the combination of transistors
112 and 125 and the combination of transistors 115 and 122 are similarly
alternately turned on and off to pass a current through the windings 136
and 137 through the points e, f, g, and h in order successively. The
direction of current flowing in the windings 134-137 is changed over
corresponding to the inversion of flux interlinkage with the windings
134-137 in accordance with the position of the permanent magnets 109 while
the position of the movable member 108 is detected by the position sensor
132, so that unidirectional thrust is generated to move the movable member
108.
When the speed of the movable member is accelerated and then the detected
speed measured by the control circuit 130 on the basis of the position
signal from the position sensor 132 reaches a certain value, the control
circuit 130 changes the transistors to be repeatedly turned on and off.
That is, on the A-phase side, the combination of the transistors 110 and
124 and the combination of the transistors 114 and 120 are respectively
replaced by the combination of the transistors 111 and 124 and the
combination of the transistors 114 and 121. On the B-phase side, the
combination of the transistors 112 and 125 and the combination of the
transistors 115 and 122 are respectively replaced by the combination of
the transistors 113 and 125 and the combination of the transistors 115 and
123. Accordingly, on the A-phase side, a current can flow only in the
winding 135, and on the B-phase side, a current can flow only in the
winding 137. In this condition, the thrust constant k.sub.f ', the induced
voltage constant k.sub.e ', the inductance L' and the resistance r' in the
motor are represented by the following equations (4):
##EQU6##
In the case where currents flow in the windings 134-137, the thrust at one
phase is represented by the equation (2) as described above. In the case
where currents flows only in the windings 135 and 137, the thrust f is
represented by the following equation (5):
##EQU7##
In the case where currents flow in the windings 134-137, the starting
thrust f.sub.m for v=0 is calculated from the equations (2) and (5):
##EQU8##
In the case where currents flows only in the windings 135 and 137, the
starting thrust f.sub.m, is calculated in the same manner.
##EQU9##
In the case where currents flow in the windings 134-137, the maximum
velocity V.sub.m for f=0 is represented by the following equation (8):
V.sub.m =E.sub.s /k.sub.e (8)
In the case where currents flow only in the windings 135 and 137, the
maximum velocity V.sub.m ' is represented by the following equation (9):
##EQU10##
FIG. 12 shows a velocity-thrust curve (a) in the case where currents flow
in the windings 134-137, and a velocity-thrust curve (b) in the case where
currents flow only in the windings 135 and 137. The resistance r' in the
case of (b) is half the resistance in the case of (a). Accordingly, the
current in the case of (b) is double the current in the case of (a).
Accordingly, the current capacity of the windings and circuit is limited
by the current capacity in the case of (a), so that the maximum thrust in
the case of (b) becomes f.sub.m /2 as shown by the dotted line in the
drawing.
As described above, the magnitude of the thrust in the case where currents
flow in the windings 134-137 and the magnitude of the thrust in the case
where currents flow only in the windings 135 and 137 are reversed
depending on the velocity. In other words, in the condition in which the
velocity is lower than V.sub.c, the thrust increases as the number of
turns of the winding increases. However, in the condition in which the
velocity is higher than V.sub.c, the thrust increases as the number of
turns decreases.
Accordingly, by control to pass currents through all the windings 134-137
in the condition in which the velocity is lower than V.sub.c and to pass
currents only through the windings 135 and 137 in the condition in which
the velocity is higher than V.sub.c, the velocity-thrust characteristics
can move on the characteristic curves (a) and (b) to attain large thrust
and high speed.
FIG. 13 shows the rising of velocity of the actuator according to the
present invention. In the drawing, the curves (a) and (b) show the rising
of velocity corresponding to the velocity-thrust characteristics (a) and
(b), respectively. In the case (a) where currents flow in all the
windings, the rising of velocity is rapid because of the large thrust, but
the maximum velocity is low. In the case (b) where currents flow in half
the windings, the rising of velocity is slow because of the small thrust,
but the maximum velocity is high. Accordingly, by switching currents from
all of the windings to half of the windings at Vc, the velocity rising
characteristics in which velocity rises rapidly and reaches its maximum
velocity rapidly can be attained as shown by the dotted-line curve of FIG.
13.
As described above, according to the second aspect of the invention, the
winding for every phase in the motor is divided into two sections and
controlled to pass currents through all the windings in the low-speed area
and to pass currents through half of the windings in the high-speed area.
Thus, an actuator or a motor having large thrust and being driven up to a
high speed can be attained.
Although this embodiment has shown the case where the winding for every
phase is divided into two sections, the invention is applicable to the
case where the winding is divided into three or more sections to more
improve the high-speed characteristics. However, in the case where the
winding for every phase is divided into a large number of | | |
