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BACKGROUND OF THE INVENTION
The present invention relates to nuclear magnetic resonance (NMR) imaging
systems and, more particularly, to a novel circuit for speeding up the
rise and fall times of current pulses utilized to generate gradient
magnetic fields in such systems.
It is now well known that NMR imaging and/or spectroscopy systems require
at least one power amplifier for each magnetic gradient field direction
utilized. These gradient power amplifiers provide the current which
generates the magnetic gradient fields, typically in the X,Y and Z
dimensions of a Cartesian coordinate system, as necessary to obtain
desired spatial resolution. Typically, the gradient power amplifiers are
modified forms of linear high-fidelity audio power amplifiers, which
typically generate current pulses in the 100-200 ampere range; the
relatively good linearity, rise times and fall times of these amplifiers
are obtained by the application of relatively high voltages and feedback
to output stages containing as many as 100 bipolar transistors. These
power amplifiers are relatively inefficient (having typical efficiencies
of less than 15%). As higher imaging speeds are utilized, greater stress
is applied to existing gradient power amplifiers, as faster rise times
require greater currents (in the same gradient coil inductance) and so
increasingly higher voltages and more power dissipation are all required.
It is therefore highly desirable to provide a current amplifier circuit,
preferably capable of being added to an NMR system between an existing
gradient power amplifier and its associated gradient coil, for providing
the faster pulse current waveform rise and fall times necessary for
higher-speed imaging use.
BRIEF SUMMARY OF THE INVENTION
In accordance with the invention, a gradient current speed-up circuit for
use in a higher-speed NMR imaging system having a gradient power amplifier
and an associated gradient coil, comprises: an energy-storage element,
having an inductance typically between 5 and 20 times the inductance of
the associated gradient coil; a plurality of semiconductor switching
elements receiving the current output of the energy-storage element; means
for connecting the associated gradient coil between selected ones of the
semiconductors devices; and means for turning the semiconductor devices on
and off in selected patterns, to cause the energy-storage element current
to be suddenly applied to and removed from flow through the associated
gradient coil.
In a presently preferred embodiment, four insulated-gate bipolar
transistors (IGBTs) are utilized in a full-bridge configuration. An
optional fifth IGBT can be utilized to abruptly remove current flow
through the associated gradient coil and reduce current fall time.
Accordingly, it is an object of the present invention to provide a novel
gradient current speed-up circuit for use in NMR imaging and spectroscopy
systems.
This and other objects of the present invention will become apparent upon
reading of the following detailed description, when considered in
conjunction with the associated drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a schematic block diagram of a single
directional-gradient-magnetic-field-providing portion of a NMR system, and
of a presently preferred embodiment of the novel circuit of invention; and
FIG. 1a is a schematic block diagram of one presently preferred embodiment
of a driver circuit for providing drive control signals to the gradient
speed-up circuit of FIG. 1.
DETAILED DESCRIPTION OF THE INVENTION
Referring initially to FIG. 1, a presently preferred embodiment of our
gradient current speed-up circuit 10 is utilized with an associated
gradient coil 11, for providing the magnetic-field gradient in one (of a
plurality) of directions within the operating volume of a NMR imaging
and/or spectroscopy system. Gradient coil 11 is normally driven by a
gradient power amplifier 12. A set of configuration switches 14 may be
utilized to either connect the gradient coil 11 between the output of
existing gradient power amplifier 12 and system common potential, or to
insert the speed-up circuit 10 between amplifier 12 and coil 11. Thus, a
first switching means 14-1 has a first selectable contact 14-1a connected
to a first selectable contact 14-2a of a second switching means 14-2, and
has a second selectable contact 14-1b connected to an input 10a of the
speed-up circuit. The first switching means common terminal 14-1c is
connected to the output of amplifier 12. The second switching means 14-2
second selectable contact 14-2b is connected to a first output 10b of the
speed-up circuit, while the second switching means common contact 14-2 c
is connected to a first end of the gradient coil 11. The remaining end of
the gradient coil is connected to the common contact 14-3c of a third
switching means 14-3, having a first selectable contact 14-3a connected to
system common potential, and a second selectable contact 14-3b connected
to a second speed-up circuit output 10c. Thus, in normal use, with the
switching means 14 configured as shown, a gradient coil current I.sub.g is
caused to flow through the gradient coil inductance L.sub.G, directly from
the output of existing system gradient power amplifier 12; the
characteristics of the gradient magnetic field thus formed is determined
by the characteristics of the gradient current I.sub.g. All three switch
means are thrown to the opposite position to connect the second selectable
contact 14-ib to the common contact 14-ic, where 1.ltoreq.i.ltoreq.3, to
utilize the gradient current speed-up circuit 10.
In accordance with the present invention, the gradient current speed-up
circuit 10 includes an energy-storing inductive element 16 having an
inductance L.sub.C which is greater than the associated gradient coil
inductance L.sub.G ; ideally, L.sub.c is between five and twenty times
L.sub.G. The presence of the high-inductance element 16 in series with
amplifier 12 output causes a substantially-constant current I.sub.c to be
sourced into a circuit node 10d, soon after the amplifier 12 is turned on.
Current I.sub.c must flow to circuit common potential, as through at least
one controllable conduction means 18. Here, a pair of paralleled
controllable-conduction means 18-1 and 18-2 are utilized. Each
controllable-conduction means 18 contains at least one controllable
solid-state device 20, and such other elements, e.g. reverse-conduction
diodes 22 and the like, as are necessary for proper operation, protection,
etc. Illustratively, each means 18 contains series-connected first and
second insulated-gate bipolar transistors (IGBT) 20-1 and 20-2, connected
between a means input 18-1a or 18-2a, and a means common potential
connection 18-1b or 18-2b. A reverse-conduction diode 22-1 or 22-2 is
connected across the collector-emitter circuit of an associated IGBT 20-1
or 20-2; diodes 22 may be integrally formed with the associated IGBT, or
may be separate diffusions or elements within a common package. The
junction between the first and second IGBTs forms an associated means
center connection 18-1c or 18-2c, at which the speed-up circuit gradient
coil output 10b or 10c is respectively taken. Associated with each
controllable-conduction device 20 is a driving means 24; a first driving
means 24-1, comprising a gate impedance 24-1a and an isolation transformer
24-1b, couples a first actuating signal V.sub.1 across the control input
(the gate-source circuit) of IGBT 20-1, and thus between terminals 18-1d
and 18-1c. Similarly, a second driving means 24-2 has a second impedance
24-2a and another transformer means 24-2b, to provide the second actuating
signal V.sub.2 between the second device input terminal 18-1e and terminal
18-1b. It will be seen that third and fourth driving means 24-3 and 24-4,
respectively, having impedance 24-3a or 24-4a and transformer means 24-3
or 24-4, respectively, transform third and fourth driving signals V.sub.3
or V.sub.4 to appear across the respective gate-source terminals 18-2d and
18-2c or 18-2e and 18-2b, of the respective controlled-conductor devices
20-1 or 20-2, of second control-conduction means 18-2.
Snubber means 26 is connected between node 10d and common potential, to
limit the high voltage generated at node l0d due to the rapid switching of
the current I.sub.C into and out of the associated gradient coil
inductance L.sub.G. Snubber means 26 includes a capacitive element 26a in
series with a dissipative element 26b, between node 10d and common
potential, and a unidirectionally-conducting diode element 26c connected
in parallel with dissipative element 26b.
Advantageously, a fifth IBGT 28 has its controlled-conduction circuit
connected from the energy-storage element 16 to circuit common potential,
i.e. with its collector electrode connected to node 10d and its source
electrode connected to common potential, and with its insulated gate
electrode connected through a current-limiting impedance 28b, to receive a
short-circuit-actuating signal V.sub.s.
Referring now particularly to FIG. 1a, the actuating signals V.sub.1
-V.sub.4 for the illustrated full-bridge speed-up circuit can be provided
with greatest versatility by use of a driving circuit 30 which utilizes
driver amplifiers 32 of number equal to the number of individual actuating
signals to be provided. Thus, here four individual driver amplifiers
32a-32d are utilized, with the output of each driver amplifier 32j, where
1.ltoreq.j.ltoreq.4, providing the actuating signal V.sub.j. Presently, we
prefer that each driver amplifier 32 be a power operational amplifier type
PA09 (available from APEX Semiconductor Co.) and the like, each receiving
a positive operating potential +V and a negative operating potential -V.
The power operational amplifier inputs receive either the signal at a
circuit control signal input 10e, or the inverse of that signal as
provided by a signal-polarity-inverting means 34. Means 34 utilizes an
operational amplifier 36, having its inverting input 36a connected to
circuit input 10e via an input resistor 38a, and connected to a feedback
resistor 38 b to the operational amplifier output 36b. The operational
amplifier non-inverting 36c is connected to common potential through a
stabilization resistor 38c. The signal at input 10e is connected to first
selectable terminals 40a--a, . . . , 40d-a of each of four switch means
40a-40d, each having a second selectable terminal 40a -b, . . . , 40d -b
receiving the inverted-polarity input signal from means 34. The common
output 40a -c, . . . , 40d -c of one of switch means 40 is connected to
the input of an associated driver amplifier 32a, . . . , 32d.
In a typical operational sequence, the control switches 40 are all
initially operated so that input 10e is connected to all of the amplifier
32 inputs; the speed-up circuit input control signal V.sub.in is enabled,
to cause each of IGBTs 20-1 and 20-2 of both of the control-conduction
means 18-1 and 18-2 to conduct. These means may be type MG100N2YS1 units
available from Toshiba and the like manufacturers. An enabling signal is
provided at the control input 12x of the power amplifier 12, to cause a
desired level (say, 25 amperes) of current I.sub.c to flow into the
energy-storage inductance 16 (configuration switches 14 having previously
been set to connect each second selectable contact 14-ib2 to the
associated common contact 14-ic, so that gradient current speed-up circuit
10 can be utilized). The substantially constant current I.sub.c flows into
node 10d and thence through the saturated devices 20-1 and 20-2 of the
paralleled modules 18-1 and 18-2, to circuit common potential. Auxiliary
short-circuiting device 28a can be operated, by suitable shorting
potential V.sub.s, to provide an additional parallel current path.
Assuming that the saturation resistance of lower devices 20-2 are
substantially equal, the potential at speed-up circuit outputs 10b and 10c
is substantially equal, so that there is substantially no potential
appearing across gradient coil 11, and there is substantially no flow of
current through the gradient coil. At the time for initiating a flow of
gradient coil current, a diametrically-opposed pair of devices 20 (as well
as device 28, if used) are driven into the open-circuit condition, with
the remaining diametrically-opposed pair of device 20 remaining in the
saturated condition; this may be accomplished by operating suitable ones
of switching means 40, which can be high-speed semiconductor devices. Now,
all of the substantially constant current I.sub.c flows through gradient
coil 11. The direction of gradient coil current flow is established by the
selection of the diametrically-opposed pair of devices 20 remaining in the
saturated condition. Thus, if control voltages V.sub.1 and V.sub.4 are
withdrawn, so that both device 20-1 in module 18-1 and device 20-2 in
module 18-2 are placed in the cut-off condition (with drive signals
V.sub.2 and V.sub.3 remaining present, so that device 20-2 in module 18-1
and device 20-1 in module 18-2 remain in the saturated condition), current
flows from node 10d, through device 20-1 of module 18-2, through output
10c, through switching means 14-3 and upwardly through the gradient coil
11, thence through switching means 14-2, into output 10b and through
device 20-2 of module 18-1, to circuit common potential; this establishes
a gradient magnetic field of slope negative to the slope which would occur
if the first and fourth driving signals V.sub.1 and V.sub.4 remain present
and the second and third driving signals V.sub.2 and V.sub.3 were removed.
The short-duration pulse of gradient current ends when the driving signals
for at least one of the parallel legs are enabled, to again provide a very
low impedance path from node 10d to circuit common potential, removing
current flow from a path through coil 11. Immediately thereafter, gradient
power amplifier 12 is disabled, to cease providing current I.sub.c. It
will be recognized that rapid change of large currents, even through the
relatively small gradient coil inductance (typically of less than 1
millihenry) can still generate relatively high voltages, which are of use
in producing desirable very short current rise and fall times and in
obtaining current pulses with very "flat" tops. Use of devices 20 having
voltage ratings of 800 or 1000 volts is preferable. It will also be
understood that, in case of switching faults, even higher voltages could
be applied to the module current input terminals 18-1a and 18-2a, and thus
across the series-connected devices 20, or across the single device 28a.
For example, if a switching fault occurs and all of devices 20/28 are
suddenly turned off, while a current I.sub.c is flowing, the node 10d
voltage V.sub.10d =L.sub.c (.DELTA.I.sub.c /.DELTA.t); snubber means 26 is
provided, and the component values selected, to limit the node 10d maximum
voltage to a value less than the maximum voltage at which any of devices
20/28a suffer damage.
While the present invention has been described with respect one presently
preferred embodiment thereof, many variations and modifications will now
become apparent to those skilled in the art. It is our intent, therefore,
to be limited only by the scope of the appending claims, and not by the
specific details and instrumentalities presented by way of description of
the presently preferred embodiment of the present application.
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