 |
Description  |
|
|
TECHNICAL FIELD
This invention relates generally to noise blankers and more particularly to
those communication devices that employ noise blankers and essentially
zero intermediate frequencies.
BACKGROUND
Those skilled in the art will appreciate the harsh operating environment of
communication devices such as mobile radios. The major contributors to a
severely noisy environment for the mobile radio include engine noise,
(both from the vehicle using the mobile radio and surrounding vehicles),
electrical interference from high power lines, and atmospheric
disturbances.
Some mobile radios have employed noise blankers to suppress or eliminate
these noise effects. The basic purpose of a noise blanker is to detect the
presence of impulse-type noise and momentarily prevent the noise in the
recovered signal from reaching the intermediate frequency (IF). For the
noise blanker to function properly, it must detect the presence of noise
and inhibit the signal path in the main receiver before the noise gets to
the point where it is to be stopped. Historically, implementation of a
noise blanker in a mobile receiver was facilitated by the commensurate
bandwidth of the main receiver and the noise blanker (i.e. each about 1
megahertz). Thus, the "race" condition was not a significant problem.
Since the bandwidths were practically the same, the delay was effectively
the same or could be compensated for by small "lump element" filters.
Modern mobile radios however, have extremely broad bandwidths. Since most
mobile radios have frequency synthesizers that can generate a wide variety
of frequencies, mobile radios today use broad bandwidth filters permitting
the mobile radio user to operate over a wide band of frequencies. Thus it
is common for a receiver to have bandwidth of 20 or 30 megahertz. However,
this bandwidth extension creates significant problems in the operation of
the noise blanker circuitry. Since the band width of the main receiver may
be twenty times the bandwidth of the noise blanker (thus making the noise
blanker delay 20 times that of the main receiver), control pulses can not
reach the blanker switch in time to prevent the noise from entering the
receiver IF. To compensate for a delay of this magnitude, a "lump-element"
filter cannot be used since the current trend is toward radio size
reduction. Hence, the size of such a filter would be prohibitive.
A solution to the delay problem was achieved using a surface acoustic wave
(SAW) filter to afford both selectivity and time delay in an appropriately
sized filter. However, SAWs are expensive commodities.
To further achieve miniaturization, microelectronic techniques are desired
in fabricating radios. Receivers producing substantially low frequency
intermediate frequency (IF) signals are known to be easier to implement
microelectronically for the intermediate stage. Since this I.F. frequency
may be substantially zero Hertz (i.e. DC or baseband), the term zero I F
(ZIF) is used in describing such an IF signal or stage. "Direct
conversion" receivers further utilizes the ZIF advantage to eliminate a
prior stage by converting an incoming signal directly to baseband. With
ZIF or direct conversion, the necessary sharp selectivity is then achieved
through lowpass rather than bandpass filtering. Since low frequency
lowpass filters are readily fabricated in monolithic form, a much greater
degree of miniaturization can be achieved in proportion to the amount of
bandpass filters being converted into lowpass.
Thus a need exists to provide effective noise blanking while
contemporaneously providing broad receiver bandwidth and radio size
reduction.
SUMMARY OF THE INVENTION
Utilization of ZIF signals in the receiver provides some advantages,
namely, it eliminates the need for complex high frequency bandpass IF
filters, and facilitates integration of the IF circuitry on an integrated
circuit (IC) chip.
Accordingly, it is an advantage of the present invention to provide noise
blanking in a "zero-I.F." receiver.
Briefly, according to the invention, an essentially zero intermediate
frequency receiver for recovering an information signal from a received
signal, which includes means for blanking noise signals which may
otherwise deteriorate performance, comprises a receiver for recovering the
information signal and a noise blanker. The receiver comprises at least
one conversion mixer for operating on the received signal to provide an
essentially baseband signal, at least one delay filter coupled to the
conversion mixer for producing a delayed essentially baseband signal, and
at least one blanker switch for operating on the delayed essentially
baseband signal to temporarily prevent recovery of the information signal
in response to a control signal. To provide the control signal, the noise
blanker is coupled to the receiver for operating on either the essentially
baseband signal or the received signal as a noise blanker input signal.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of a radio employing a noise blanker of the
present invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
Referring to FIG. 1, a noise blanker 28 of the present invention is
included in a portion of an FM receiver 100 having a main receiver 10.
Preferably, the circuits comprising this portion of the receiver 100
utilize bipolar and metal oxide semiconductor (BIMOS) technology for
integrating the circuit on an IC chip. The receiver 100 may be used in
radio communication units, such as mobile two-way transceivers.
In the receiver 100, a received radio frequency (RF) signal or intermediate
frequency (IF) 110 is amplified by a preamplifier 12, which produces an
amplified signal 115. The input signal 115 from the preamplifier 12 is
supplied to each of two parallel, substantially identical paths
32A-40A-S1-33A and 32B-40B-S2-33B.
Conventionally, elements 32A and 32B are down-conversion mixers that
translate the incoming signal 115 to essentially baseband. A
down-conversion frequency is supplied in quadrature to both mixers 32A and
32B using a phase shifter 36 or equivalent to provide two signals in phase
quadrature. The frequency of the LO signal 112 is selected such that it is
substantially equal to the frequency of the received signal 110. In more
detail, the phase-shifting circuit 36 receives the local oscillator
(f.sub.DOWN) waveform 112 and produces an inphase waveform (I) and a
quadrature waveform (Q) in response to the f.sub.DOWN waveform. The down
mixers 32A-B convert the signal from the RF to essentially baseband
frequency. Therefore, the pair of IF signals 125A and 125B have a
substantially low frequency and are modulated at the baseband frequency.
The outputs of mixers 32A and 32B are fed to two identical low pass
filters 33A and 33B which remove any received spurious signals and limits
the noise bandwidth of the receiver 100.
The respective outputs of these lowpass filters are coupled to a
demodulator 52. Subsequently, a modulating signal may be recovered by any
suitable demodulation technique at the demodulated output. The
demodulation technique may preferably comprise upmixing, by an upmixer,
the ZIF signal with a second high frequency local oscillator, and applying
the output of the upmixer to a well known phase lock loop (PLL) or other
type demodulator. In the preferred embodiment of the invention, modulating
signal recovery is achieved by applying the output of each of the low pass
filters 33A and 33B to a pair of suitable up conversion mixers, which
produces a pair of upmixer signals in phase quadrature. The up mixers thus
convert the baseband signals up to a convenient frequency for further
processing and demodulation.
According to the present invention, the conventional zero intermediate
frequency (IF) receiver has been modified by adding delay low pass filters
40A and 40B at the output of the downmixers 32A and 32B, series switches
S1-S2 at the inputs of the baseband bandpass or lowpass filters 33A and
33B, and shunt switches S3 and S4 from the inputs of the lowpass filters
33A and 33B to analog ground. In addition, the noise blanker 28 controls
the selective opening and closing of the switches S1-S4 in the main
receiver 10.
After down mixing in the down mixers 32A and 32B, the pair of substantially
zero baseband signals 125A and 125B are coupled to the pair of delay
elements 40A and 40B. The pair of delay elements 40A and 40B may be
implemented as a simple lowpass filter using a distributed RC delay line
or with discrete components (resistors and capacitors). The lowpass
filters 40A and 40B time delay the substantially zero IF signal for
approximately 3 micro seconds and thus provide the major amount of time
delay in the main receiver 10.
The noise blanking switches S1-S4 provide the means by which the received
signal is interrupted and thus prevented from entering the pair of lowpass
filters 33A and 33B. The blanker switches S1 thru S4 may be implemented
using any suitable technology and may be, for example, one or more field
effect transistors (FET's) configured either in series and/or in shunt (to
the received signal path) to provide the required attenuation. The blanker
switches S1-S4 are positioned between the delayed filter 40A and 40B and
the main selectivity (the lowpass filters 33A and 33B) so that the main
receiver 10 may "blank" after the downmixers 32A and 32B.
Normally, the blanker switches S1-S2 are "closed" and S3-S4 are "opened" to
couple the output of the downmixers 32A and 32B to the input of the pair
of the lowpass filters 33A and 33B to allow the received signal to be
processed by the demodulator 52 and subsequent circuitry.
Thus, when the noise blanker 28 determines that a noise condition exists,
the blanker switches S1-S2 are momentarily "opened" (by asserting a
control input 58) to prevent the received signal from entering the lowpass
filters 33A and 33B and being demodulated by the demodulator 52. In
addition, the shunt switches S3 and S4 connect the delay filters' outputs
to analog ground when noise is present to prevent glitches. In this
manner, a long recovery time is prevented in the delay filter output
circuits (which act as a current sink) which may take place if the delay
filters' outputs are allowed to float to their maximum or minimum voltage
levels. The "open" duration is appropriately set to prevent the recovered
signal containing the noise from entering the lowpass filters 33A and 33B,
after which the blanker switches S1-S2 "close" and the blanker switches
S3-S4 revert to an "open" position permitting normal operation.
To provide the control signal 58 to control the switches S1-S4, the noise
blanker 28 including filters 60 and 70 is coupled to the main receiver 10
for operating on either the essentially baseband signal 125A or 125B or
the received signal 115 as a noise blanker input signal 128. The filter 60
sets the bandwidth of the noise blanker 28 and determines the amount of
frequency spectrum that the noise blanker 28 will monitor for noise.
Depending on how the noise blanker 28 is connected to the main receiver 10
to determine what the noise blanker input signal 128 is, the filters 60
and 70 are either bandpass or lowpass filters. The filtering is greatly
simplified from a bandpass filter centered at the noise blanker RF
frequency to a bandpass filter centered at DC which becomes a lowpass
filter.
In a fixed RF channel embodiment, independent of the desired RF frequency,
the noise blanker 28 accepts the received signal 115 at an RF bandpass
filter 60 tuned to a fixed RF channel where noise is expected. Since the
bandwidth of the main receiver 10 is broad there may be several mobile
radio users transmitting in the allotted spectrum. Thus the tuned RF
bandpass filter 60 of the noise blanker 28 must be set or tuned to monitor
a portion of the frequency band that is not being used by other carriers
or information signals since they may be interpreted as noise and the main
receiver 10 will be inhibited. The bandpass filter may be implemented by
any topology that facilitates tuning and may be for example, a 3
pole-coupled resonator filter having a 1 megahertz bandwidth or suitable
equivalent.
On the other hand, in the preferred embodiment for easier microelectronic
implementation, the noise blanker 28 accepts one of the essentially
baseband signals 125A or 125B or a weighted sum of each at the more
desired lowpass filter 60. In this embodiment, the noise blanker 28 has an
RF channel centered at the desired receive baseband frequency since the I
and Q signals are always centered at baseband. Hence, the filter 60 may be
implemented easily as a lowpass filter having a bandwidth of approximately
0.5 megahertz using resistors and capacitors, as opposed to high loaded Q
band pass filters or SAW delay lines. As the IF frequency drops and
approaches zero, this embodiment is preferred to enable usage of more
lowpass filters.
With either embodiments, the band-limited noise signal is then applied to
an automatic gain controllable (AGC) amplifier 64 which accepts an AGC
input signal at terminal 68. The AGC signal applied at port 68 of the
amplifier 64 increases or decreases the gain of the amplifier 64 in the
well known AGC operation.
The now appropriately amplified noise signal is applied to a tuned RF
bandpass filter 70 in the fixed RF channel embodiment or a simple lowpass
filter 70 in the preferred embodiment to again band-limit the signal which
is then coupled to a pulse detector 72. The pulse detector 72 monitors the
amplified band-limited signal and compares it to a predetermined threshold
to determine when noise spikes (or pulses) are present. When the noise
peaks exceed the predetermined threshold the pulse detectors 72 outputs a
pulse indicating that excessive noise is present. The pulse output from
the pulse detector 72 is amplified in an optional separate pulse amplifier
74 (or incorporated in a pulse shaper 76) which provides sufficient gain
to the pulse to trigger a pulse shaper 76.
The pulse shaper 76 accepts the amplified "trigger" pulse and first
generates a substantially rectangular pulse which is then shaped into a
trapezoidal shape or any other desirable shapes have sloped rising and
falling edges and having a predetermined pulse duration. The duration of
the pulse or the control signal 58 generated by the pulse shaper 76, is
set to allow sufficient time for the blanker switches S1-S4 to reach and
maintain maximum attenuation, thus preventing the noise signal, being
delayed by the pair of delayed filters 40A and 40B, from entering the pair
of lowpass filters 33A and 33B. Accordingly, the duration of the pulse
generated by the pulse shaper 76 may be set to an appropriate duration to
allow the blanker switches S1-S4 to reach full attenuation and remain
"open" until the noise signal has sufficient time to pass through the
delay filters 40A and 40B taking into account the varying parameters.
A rate shutoff circuit 86 is shown as an optional feature for the noise
blanker 28. As is known, rate shutoff circuit measure the repetition rate
of detected noise pulses without regard to their amplitude. If the rate
exceeds a predetermined value, the circuit 86 will disconnect the blanking
function from the essentially Zero IF signal, since if the repetition is
too high, no signal will be recovered anyway since "blanking" will be
continuous.
As previously mentioned, the amplifier 64, and thus the noise blanker 28,
is controlled by the AGC signal. Generally, an AGC signal is commonly used
in AM receivers as a control for amplifiers. Basically, the goal of the
AGC circuit is to reduce the gain of the blanker RF channel when the
desired signal increases, thereby desensitizing the blanker 28 and
increasing the minimum noise pulse amplitude required to initiate
blanking. As the desired signal level increases, the smaller noise pulses
no longer create objectionable interference, whereas blanking would create
interference. Accordingly, an AGC circuit including an AGC RF amplifier
and detector 78 controls the gain of an AGC amplifier 64 and reduces the
gain to reduce the sensitivity of the noise blanker 28 when the desired
received signal exceeds the threshold level.
In the preferred embodiment of using a lowpass filter 60 in the noise
blanker 28 to feed in the essentially baseband signal 125A or 125B, both
noise and the desired signal is received since the blanker's RF channel
(60) is centered at the desired receive (essentially baseband) frequency.
Accordingly, the noise signal and the desired signal received and filtered
by the lowpass filter 70 is utilized as a feed back signal 145 to control
the AGC amplifier and detector 78. On the other hand, in the alternate
embodiment of the fixed RF channel being tuned for expected noise in the
noise blanker 28, the desired signal is not present in the fixed RF
channel since the tuned RF filters are intentionally tuned to eliminate
the desired signal. Therefore, another source of AGC control is needed.
Hence, in the main radio receiver 10 (maybe from the demodulator 52), a
received signal strength indicator (RSSI) signal from the RSSI 160 may be
utilized to indicate the strength of the signal (including noise)
received. Thus, the AGC signal can be developed from the RSSI 160 where
the RSSI signal output from the RSSI 160 is a DC voltage which varies
proportionally to the signal strength of the received signal including
noise. Coupled from the RSSI 160, the RSSI signal is applied to the AGC
port 68 of the amplifier 60 to control the gain in the well known AGC
operation.
In summary, the noise blanking circuitry can be greatly simplified for a
receiver with a zero IF because the IF filtering is done at audio
frequencies. In this case, filtering normally done with narrow-band RF
tuned bandpass circuits or the equivalent can be replaced with lowpass
filters. The filtering is greatly simplified from a bandpass filter
centered at the noise blanker RF frequency to a lowpass filter centered at
DC. The lowpass filter can be implemented using either simple resistors
and capacitors (RC) inductors and capacitors (LC), or active integrated
filters. If lossiness is not a big problem, RC's are probably preferable
since they are easier to integrate.
* * * * *
|
|
 |
|
 |
Description |
|
