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BACKGROUND OF THE INVENTION
1. Field of the Invention
This invention relates to the communication of signals, and more
particularly to the transmission and reception of digital signals.
2. Background of the Related Art
Communication systems are used to transmit and receive signals over a
communication channel. One common form of communication system is a
telephone system for transmitting and receiving voice and data signals.
FIG. 1 is a pictorial representation of a telephone communication system
10. Central office 11 sends telephone signals over wires 12 to central
office 14. Similarly, central office 14 can send signals to central office
11 over wires 12. Wires 12 are supported by poles 16.
The telephone communication system 10 is preferably a digital communication
system; i.e., digital signals representing digital data are sent over
wires 12. When carrying digital data, these wires 12 are often referred to
as "T1" lines. This digital data can represent voice and other types of
analog data, as well as purely digital information. Since the distances
between offices 11 and 14 may be relatively large, the digital signal
carried by wires 12 may pick up "glitches" Or "noise". To reduce this
noise, several repeater apparatus 18 are preferably placed on intermediate
poles 16 or at the receiving offices 11 and 14. A repeater apparatus 18
detects and then retransmits the digital signal so that the original
digital values of the signal are not obscured by noise. Using several
repeater apparatus 18, a signal may be transmitted a long distance over
wires 12 and arrive at a receiver including virtually all of its original
information
A repeating apparatus 18 and a receiving apparatus 19 are shown in FIG. 2.
Receiving apparatus 19 will typically be provided at receiving offices 11
and 14. Clock and data recovery apparatus 20 receives a signal from T1
line 12 and derives a clock signal therefrom. This can be accomplished,
for example, by oversampling, as is well known to those skilled in the
art. The data signal output of apparatus 20 retains the information
contained in the original signal on T1 line 12. A jitter attenuator 22 is
coupled to the clock and data outputs of the apparatus 20 to remove
digital "jitter".
Jitter is defined as intermittent phase shifts of a digital signal. An
example of jitter is shown in FIG. 3. Waveform 23 is an example of an
ideal, original waveform transmitted from a source. Waveform 24 is a
waveform that includes jitter. Jitter shifts 26 can be caused by
repeaters, other equipment, inductive delays, etc. Jitter can accumulate
over time and eventually cause the repeated waveform to be an entire pulse
width off from the original waveform. Thus, over many repeaters, jitter
can cause uncertainty in which time period a pulse occurs. Timing and
information errors at the receiving end can result from this effect.
Problems from jitter can also occur in the clock and data recovery
apparatus 20 shown in FIG. 2. The phase locked loop typically used in
apparatus 20 can oscillate for a short time at a specific frequency
without receiving a timing reference from a pulse, but will begin to drift
and produce an incorrect clock signal if there is a long interval of zeros
in a signal, i.e. no pulse edges. Such an interval of zeros can also cause
jitter.
The problem of jitter has been addressed in the prior art by an analog
jitter attenuator. Typically, an analog jitter attenuator 22 is connected
to the clock and recovery block 20 to receive the clock and data signals.
The jitter attenuator produces an output clock at an average of the input
clock frequency. Since the output clock is the average of the input clock
frequency, short term variations in the input signal are filtered out.
These analog jitter attenuators 22 of the prior art have many of the
accuracy, stability and noise problems associated with analog devices.
An article, "Digital Phase-Locked Loop with Jitter Bounded", by Stephen M.
Waiters and Terry Troudet (IEEE Transactions on Circuits and Systems, July
1989), describes a digital jitter attenuator that uses a digital
phase-locked loop to generate signals that satisfy pre-imposed
requirements on jitter over a given range of frequencies. Control of the
jitter is obtained by means of a frequency-phase window comparator which
compares a bit overflow/underflow of a digital controlled oscillator to a
fixed frequency-phase window.
A problem with this prior art digital jitter attenuator occurs due to the
high frequency clock required in the numerically controlled oscillator
(NCO) to produce a jitter-free data signal. A high frequency clock is
required to achieve an accurate output clock that can be finely adjusted.
In the prior art, a large number of bits is required in the NCO to achieve
such a high resolution. However, it is difficult to process (i.e., add) a
large-bit number in the short period of the high frequency clock. The
prior art is thus limited in the frequency resolution of the signal clock
it can output and the amount of jitter it can reduce.
What is needed is an apparatus and method that will reduce jitter in
transmission systems using an accurate, high resolution digital system.
SUMMARY OF INVENTION
The present invention addresses the problems in the prior art by providing
a method and apparatus for the reduction of jitter in digital
communication systems. A digital apparatus is provided that uses a
numerically controlled oscillator (NCO) that preferably includes a low
frequency portion and a high frequency portion to output a compensation
clock. The compensation clock controls a buffer to output a digital signal
substantially free of jitter.
In the present invention, a jittered signal is input to a jitter sensing
mechanism that produces a digital signal in accordance with the amount of
jitter on the input signal. A clock generator mechanism, including a low
frequency portion and a high frequency portion, receives the digital
signal. The high frequency portion outputs a clock of a compensation
frequency determined, in part, by the digital signal from the jitter
sensing mechanism and sends the compensation clock to a buffer which
produces an output signal with reduced jitter.
The jitter sensing mechanism preferably includes a input counter and an
output counter that are intentionally out of phase of each other. An input
clock is derived from the input signal and and is coupled to the input
counter. As the input counter value arrives at a predetermined maximum
count, the output counter value is latched to the address bus of a ROM
containing a look-up table. If the output counter value is not a
predetermined number of degrees out of phase with the input counter value,
a timing discrepancy is present on the input clock, possibly caused by
jitter. A correction coefficient is read from the look-up table of the ROM
to control an NCO.
The NCO of the present invention is preferably divided into two portions. A
low frequency portion includes an adder and a latch. The adder receives
the coefficient signal from the look-up table of the ROM and continuously
adds the coefficient value to itself. Each sum is output to a latch, which
sends the sum back to the adder according to a clock provided by the high
frequency portion. The carry out (CO) signal from each sum is preferably
output from the adder to a high frequency portion of the NCO.
The high frequency portion preferably includes a high frequency clock that
is divided to a lower frequency and input to the latch of the low
frequency portion. A frequency divider is enabled by the CO from the low
frequency portion that has been passed through an edge detect circuit. The
edge detect circuit preferably enables and disables the divider to
compensate for jitter. A compensation clock signal is output from the
divider and is used to provide a jitter-free data output signal using data
stored in the buffer.
These and other advantages of the present invention will become apparent to
those skilled in the art after reading the following descriptions and
studying the various figures of the drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a pictorial representation of a communication system used for the
present invention;
FIG. 2 is a block diagram of a prior art repeater, clock and data recovery
apparatus and jitter attenuator;
FIG. 3 is a diagram of an original signal and the signal including jitter;
FIG. 4 is a block diagram of the jitter attenuator of the present
invention;
FIG. 4a is an illustration of the input and output counter values used in
the present invention;
FIG. 4b is a preferred ROM-based look-up table used in the present
invention;
FIG. 4c is a preferred ROM-based look-up table used in an alternate
embodiment of the present invention; and
FIG. 5 is a block diagram of a numerically controlled oscillator of the
present invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIGS. 1, 2 and 3 were discussed in the Background section of this
specification. FIG. 4 is a block diagram of a jitter attenuator 30 in
accordance with the present invention which replaces the prior art analog
Jitter attenuator 22 illustrated in FIG. 2. Jitter attenuator 30
preferably includes an input counter 32, an output counter 34, an AND gate
36, a buffer 38, a phase detect latch 40, a read only memory (ROM) 42, and
a numerically controlled oscillator (NCO) 44. Input counter 32 is coupled
to an input clock signal line 46 which provides an input clock from a
clock and data recovery apparatus 20 (FIG. 2 ). The input counter 32
counts with each pulse of the input clock. By way of example, a suitable
counter is a 5 bit modulo 32 counter. The output of the input counter 32,
in this example, is a 5 bit number provided on bus 48.
Output counter 34 can be the same type of counter as input counter 32. The
count of output counter 34 is set to a predetermined phase difference from
input counter 32. In the preferred embodiment, the count of output counter
34 is set to be 180 degrees out-of-phase with the count of input counter
32, explained in further detail with reference to FIG. 4a. Output counter
34 is coupled to buffer 38 by 5-line bus 50.
Buffer 38 is preferably a RAM-based FIFO ring (i.e. random access memory
(RAM) based) buffer, such as a 32 bit dual port RAM. Other kinds of
buffers can also be used. Buffer 38 preferably includes a write address
input WA which is coupled to input counter 32 by bus 48. A read address RA
input of buffer 38 is preferably coupled to output counter 34 by bus 50,
and a data input port DI of buffer 38 is preferably coupled to input data
line 52 from a clock and data recovery apparatus 20 (FIG. 2). The data
output port DO of buffer 38 is coupled to an output line 54 which carries
a de-jittered digital output signal for further processing in the central
office. Buffer 38 thus functions as an output mechanism for the jitter
attenuator 30.
The AND gate 36 is coupled to each line of bus 48. Therefore, in this
example, this is a five input AND gate. The output of AND gate 36 is
coupled to a latch enable port LE of phase detection latch 40 by a line
37. A suitable phase detection latch 40 includes a S flip-flops, each of
which store one bit of data. The data input bus DI of latch 40 is coupled
to bus 50 of output counter 34. The data output bus DO 56 from latch 40 is
coupled to ROM 42.
The input address ports IA of ROM 42 are coupled to the output data bus of
latch 40 by bus 56. By way of example, ROM 42 may take the form of a
32.times.13 ROM. The output data ports DO of ROM 42 are coupled to NCO 44
by bus 58, which, in the described embodiment, takes the form of a 13 bit
bus. NCO 44 also is preferably coupled to a high frequency clock line 60.
NCO 44 is a clock generator mechanism that outputs a compensated clock on
line 62, which is coupled to an input port of output counter 34. NCO 44 is
described in detail below with reference to FIG. 5.
In an alternate embodiment, the components of jitter attenuator 30 are made
by constructing an application-specific integrated circuit (ASIC). Digital
components such as counters, ROM, flip flops, etc. can be designed in an
ASIC using logic gates, etc., and is well known to those skilled in the
art.
The operation of the preferred embodiment of jitter attenuator 30 is
briefly described as follows. The input clock on line 46 and input data on
line 52 are provided by a prior art clock and data recovery apparatus 20.
Input data on line 52 is sent to the DI port of buffer 38. The input clock
on line 46 increments the value of input counter 32. In the described
embodiment, input counter 32 is initialized at a count of 16, which is
half the preferred maximum count of 31, as illustrated in FIG. 4a. Output
counter 34 is initialized at a value of 0, which is 180 degrees
out-of-phase from the initial value of the input counter, as is also
illustrated in FIG. 4a. The input counter 32 value is output on bus 48 and
is sent to the RA port of buffer 38. Buffer 38 uses the input counter
value as a memory address in the buffer's RAM and stores the input data on
line 52 at that address.
The inputs of AND gate 36 are coupled to bus 48. If all N bits (in the
preferred embodiment, 5 bits) are high (1's), a terminal or predetermined
maximum count of input counter 32 has been reached and a high signal is
output from AND gate 36. Preferably, this terminal count is 31 on the
5-bit bus 48. The high signal output of AND gate 36 is coupled to the LE
input of phase detection latch 40 by line 37.
Output counter 34 preferably starts counting at a value of 0. Output
counter 34 increments its counter value with each clock pulse on clock
line 62. The output counter 34 outputs a value on bus 50 which is coupled
to the RA port of buffer 38. The output counter value is used to address a
memory location in the RAM of the buffer and data stored at that memory
location is output on line 54. Data is output from the buffer 38
preferably after the buffer is about half full; this occurs when the input
counter 32 has reached a maximum count of 31 in the present example.
The output counter value is also coupled to phase detection latch 40 by a
bus 50. Phase detection latch 40 is operative to latch onto the data on
bus 50 when line 37 provides a latch enable signal, i.e. when input
counter 32 is at a predetermined maximum counter value. When the input
counter outputs the maximum value of 31, the output counter should ideally
output a value of 15, which is 180 degrees out-of-phase with the input
counter value. The latched output counter value is coupled to the Input
Address IA port of ROM 42, where a coefficient value stored in a look-up
table is addressed.
A preferred look-up table 66 stored in ROM 42 is shown in FIG. 4b. The
address shown in the left column is the latched output counter value
stored in latch 40. Since the output counter value is 180 degrees
out-of-phase with the input counter value in this preferred embodiment,
the output counter value should be about 14 or 15. Addresses 14 and 15
have a coefficient value of 188A hex stored at that location in memory;
this coefficient is output on bus 58 to NCO 44. NCO 44 uses the
coefficient to determine the frequency of the compensation clock on line
62. NCO 44 drives the output compensation clock in a frequency range
centered preferably at 1.544 MHz, which is the frequency standard used for
T1 transmission lines. Of course, other frequencies can be output from the
NCO 44, such as 2.048 MHz for the European standard E1.
If the output counter value is not 14 or 15, then the output and input
counters are no longer 180 degrees out-of-phase. This indicates that there
is a timing discrepancy between the input counter and the output counter.
If a timing discrepancy occurs, a different coefficient is output by ROM
42 to adjust the frequency of the compensation clock developed by NCO 44.
The addresses input to ROM 42 are, for convenience, paired. Each pair of
addresses is assigned a coefficient in the look-up table of ROM 42.
Preferably, each address pair above the 14-15 pair produces a coefficient
which decreases the NCO output clock frequency by about 44 Hz, and each
address pair below the 14-15 pair produces a coefficient which increases
the NCO clock frequency by about 44 Hz. For example, if the latched output
counter value on bus 56 is 17, it means that the input counter is counting
slightly slower than the output counter. In other words, by the time the
input counter counted to 31, the output counter had counted to 17. The
output counter value 17 is addressed in ROM 42, and a coefficient of 1899
hex from the look-up table of FIG. 4b is output from the ROM on bus 58 to
NCO 44. NCO 44 receives the coefficient, which tells it to slightly
decrease the frequency of the compensation clock by about 44 Hz on line 62
which drives output counter 34, i.e. the output counter is slowed to match
the slower input counter. Output counter 34 is thus triggered to output a
count value at a slightly slower speed to align itself with the input
counter value; this slowed output counter value is received at buffer 38
and the data signal at that address is output on line 54 at a compensated
clock frequency to produce a jitter-free signal. As mentioned previously,
the buffer 38 is ideally half full, with 16 of 32 locations filled.
Jittered data input on line 52 may "fill up" some of the extra buffer
memory locations faster than the OUT counter outputs the data on line 54;
or the OUT counter may output data faster than jittered data is being
input. The buffer 38 serves to "absorb" jittered data so that the OUT
counter can output jitter-free data from the buffer at the compensated
clock rate.
In circuit 30, the input counter 32, output counter 34, AND gate 36, phase
detection latch 40, and ROM 42 can collectively be considered a jitter
sensing mechanism, since they function to provide a coefficient to NCO 44
that adjusts the compensation clock to compensate for jitter.
Using circuit 30, a jittered input dock is corrected by slowly increasing
and decreasing the frequency of a derived clock to compensate for the
phase shift of the jitter. The output of the circuit 30 is therefore
substantially jitter-free. Since, in the described embodiment, a
coefficient is sent to the NCO 44 from ROM 42 only when the input counter
reaches a count of 31, the compensation clock signal on line 62 is
adjusted every 32nd pulse of the input clock. In practice, this proves to
be quite effective in substantially removing jitter from a digital signal.
FIG. 4c shows an alternate embodiment of a look-up table 66' stored in ROM
42. In table 66', many addresses in the ROM are assigned unique
coefficients. This arrangement provides greater resolution and thus a more
accurate frequency for the compensation clock output from NCO 44 than the
"bracketed" table shown in FIG. 4b, where two adjacent addresses are
assigned the same coefficient. In addition, if the input signal is
temporarily lost, the look-up table of FIG. 4c will allow for a faster
convergence than the table of FIG. 4b. Look-up table 66'is nonlinear;
i.e., several addresses at each end of the range (0-7 and 24-31) are
associated with one coefficient value. This nonlinearity quickly forces
the jitter attenuator to the middle range of addresses (8-23) which are
optimized for systems with a relatively minor amount of jitter on their
input data line. If, however, a large amount of jitter occurs on the input
signal so that the output counter varies, for example, within 90 degrees
of the input counter (i.e., the output counter value is, for example, 5 or
27 when the input counter value is 31 ), then the look-up table of FIG. 4c
is not as efficient in de-jittering the signal as is the look-up table of
FIG. 4b. Thus, in the case of extensive jitter, the linear look-up table
66 of FIG. 4b may produce better results.
FIG. 5 is a block diagram of the NCO 44 of the present invention. The NCO
44 includes a low frequency portion 67 and a high frequency portion 72.
The low frequency portion preferably includes an adder 68 and a latch 70.
The adder 68 can be, for example, a 13-bit adder at a port A. Adder 68
receives a coefficient from ROM 42 on bus 58, which is a 13-line bus in
the described embodiment. Adder 68 is operative to add the inputs at A and
B and output the sums on bus 74, which is also a 13-line bus in this
embodiment. The adder also outputs the carry out (CO) bit from the adding
operation on line 75, which is coupled to the high frequency section 72.
The data input port (DI) of latch 70 is coupled to bus 74, and the output
Q of latch 70 is fed back to input B of adder 68. The clock input 76 of
latch 70 is coupled to high frequency section 72. A suitable latch that
can be used for latch 70 include 13 D flip flops.
A high frequency portion 72 in accordance with the present invention
includes a high frequency clock on line 60, a divider 82, and an edge
detector circuit 84. In this described embodiment, the high frequency
clock has a frequency of about 100 MHz. The high frequency clock is input
to frequency divider 82, which is preferably a counter that divides the
frequency of the clock signal; in the described embodiment, divider 82 is
a 6-bit counter that divides the 100 MHz clock frequency by 64, producing
a 1.56 MHz clock on the output 62. Of course, other dividing factors can
be used, such as a divide by 48 to produce a 2.083 MHz clock (later
adjusted for the E1 standard). The most significant bit (MSB) from divider
82 is used for the output (compensation) clock on line 62. The inverse of
the MSB value is coupled to the enable input E of latch 70 by line 76.
The high frequency clock on line 60 also is input to edge detector circuit
84. Edge detector circuit includes latch 88, latch 90, and NAND gate 92.
The high frequency clock on line 60 is coupled to the enable E inputs of
both latches 88 and 90. Latch 88 receives the CO from adder 68 on line 75
at its D input. Latch 90 receives the output Q from latch 88 at its D
input. The two inputs of a NAND gate 92 are coupled to the Q outputs of
latches 88 and 90. The output of NAND gate 92 is developed on line 94.
Line 94 is coupled to the Count Enable of divider 82 and is operative to
disable and enable the divider 82.
The circuit illustrated in FIG. 5 operates as follows. A coefficient from
the look-up table in ROM 42 is input as a digital signal on bus 58 to the
A input of adder 68. The coefficients in the look-up table of ROM 42 are
selected to provide a CO from adder 68 after a certain number of
summations to generate a specific clock frequency.
A preferred method to calculate coefficients as shown in FIGS. 4b and 4c is
described as follows. A coefficient is defined as the frequency of the
high frequency clock (on line 60) divided by the output frequency. For
example, a high frequency clock at 100 MHz and an output frequency of
1.544 MHz produces a coefficient of 100 MHz/1.544 MHz=64.76684. The high
frequency portion 72 of the NCO divides by 64, so it is the fractional
part of the coefficient that is handled by the low frequency portion 67.
To convert the fractional part (0.76684) to an NCO coefficient, the
fractional part is multiplied by 2.sup.N, where N is the number of bits in
the adder. For example, using a 13 bit adder, the NCO coefficient in the
example is 0.76684.times.2.sup.13 =6282 =188A hex. This is the center
value in the look up table and corresponds to the output counter being in
synchronization with the input counter. To compute the remaining values in
the look up table, a frequency tolerance in the communication system and
jitter attenuator is used. In a typical digital telephone system, for
example, the jitter has a maximum range of about 138 Hz peak-to-peak, or
+-69 Hz. The tolerance of the input clock being sent into the jitter
attenuator is about +-200 Hz. A tolerance for the crystals of the counters
used in the preferred embodiment increases the total tolerance to about
+-300 Hz. If the entire range of 600 Hz is divided by the number of
entries in the look up table (16 in the described embodiment), a step size
of 37.5 Hz results. This step size is added to or subtracted from the
output frequency successively to compute the other coefficients. To make
the step size an even hexidecimal number, an offset of F (hex) is chosen
in the described embodiment (about 44 Hz between coefficients).
Adder 68 adds the coefficient at input A to the value at input B, which is
the immediately previous sum that was output from the adder; the new sum
is then output on bus 74. The CO from this sum is a "low frequency" signal
(relative to the high frequency clock on line 60) output on line 75.
The high frequency clock on line 60 is input to divider 82 and edge
detector circuit 84. Divider 82 is preferably a 6-bit counter that counts
to 64, incrementing with each high frequency clock pulse. The MSB of the
divider output is the output compensation clock signal on line 62. In this
described embodiment, the compensation clock on line 62 from divider 82 is
thus 100 MHz/64=1.5625 MHz. The edge detector circuit 84 receives the high
frequency (100 MHz) clock at the clock input of latch 88 and outputs a low
signal on line 94 when a CO signal is detected from adder 68 at the data
input of latch 88. The low signal on line 94 deactivates the Count Enable
on divider 82, thus halting the count of divider 82 for one period of the
high frequency clock. The edge detector circuit thus functions to slow the
divider 82 relative to the high frequency clock signal and divide the high
frequency clock by one more count, i.e., in the described embodiment, the
high frequency clock is divided by 65 instead of 64. The compensation
clock is thus 100 MHz/65=1.538 MHz for that particular cycle of the high
frequency clock.
The MSB output of divider 82 is also inverted (for timing reasons) and
input to the enable input of latch 70 on line 76. Latch 70 thus allows the
sum from adder 68 to feed back to the adder at the clock rate of the
compensation clock on line 62. The summing of the coefficient provides a
carry out (CO) on line 75 at a predetermined frequency at the D input of
the edge detector circuit 84. The CO from adder 68 determines when edge
detector circuit 84 deactivates divider 82 and thus controls the fine
tuning of the divider 82 to provide an adjusted compensation clock. For
example, the desired frequency for a T1 transmission line is 1.544 MHz. If
edge detector circuit 84 has not been activated with a CO pulse, a clock
with a slightly higher frequency is output from divider 82. If the edge
detect circuit 84 is activated, a clock with a slightly lower frequency is
output from divider 82. The frequency of the compensation clock is thus
adjusted either higher or lower than the desired clock frequency of 1.544
MHz to compensate for the timing errors on the input signal. The NCO
tracks the average input signal frequency on line 46 using the coefficents
from the look-up table of ROM 42, the buffer 38 absorbs the jitter and the
jitter on output signal line 54 is substantially reduced.
The present invention includes a high frequency portion that only must
increment a 6 bit counter in the period of the high frequency clock
instead of having to add large-bit numbers in an adder, such as adder 68.
This allows a much higher clock frequency to be used and thus a higher
resolution in output clock frequency. In order to meet the AT&T 62411
standard jitter specification of 0.02 uI (unit intervals), a high
frequency clock must be used. For example, in the T1 standard of 1.544
Mbits/sec, a unit interval is 1/1.544 MHz or about 648 nanoseconds; a 0.02
uI specification of intrinsic jitter thus allows a minimum of a 12.96
nanosecond (about 77 MHz) clock. A 100 MHz clock with a 10 ns period will
provide the desired resolution to meet this specification, and the high
frequency portion of the present invention allows this to be accomplished.
While this invention has been described in terms of several preferred
embodiments, it is contemplated that alterations, modifications and
permutations thereof will become apparent to those skilled in the art upon
a reading of the specification and study of the drawings. It is intended
that the claims include all such alterations, modifications and
permutations as fall within the spirit and scope of the present invention.
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