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Claims  |
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What is claimed is:
1. A method for communicating comprising the steps of:
transmitting a repeated codeword preamble code from a first station to a second station, said repeated codeword preamble code comprising a repeated series of the same subcode;
receiving said repeated codeword preamble code at said second station;
generating a plurality of correlation spikes in response to receiving said repeated codeword preamble code at said second station, each of said correlation spikes corresponding to a separate appearance of said subcode at said second station;
determining receipt of said repeated codeword preamble code upon simultaneous detection of more than one of said correlation spikes;
synchronizing a receiver at said second station in response to said repeated codeword preamble code;
transmitting a data message from said first station to said second station; and
receiving said data message at said second station.
2. The method of claim 1 wherein said step of receiving said repeated codeword preamble code at said second station comprises the step of detecting said repeated codeword preamble code by correlating to said subcode and generating a plurality of
correlation spikes thereby, one correlation spike for each occurrence of said subcode in said repeated codeword preamble code.
3. The method of claim 2 wherein said step of detecting said repeated codeword preamble code comprises the steps of
detecting a predefined number of said plurality of correlation spikes, and
performing a verification of received data integrity of a data message following said repeated codeword preamble code.
4. The method of claim 2 wherein said step of correlating to said subcode comprises the step of inputting said repeated codeword preamble code to a matched filter configured to detect said subcode.
5. The method of claim 2 wherein said step of correlating to said subcode comprises the step of inputting said repeated codeword preamble code to a mismatched filter configured to detect said subcode.
6. The method of claim 1 wherein said subcode comprises a member of a code family selected from the group of code families consisting of: Barker codes, Boztas codes, minimum peak sidelobe codes, Kasami codes, Gold codes, and Matsufuji codes.
7. The method of claim 1 wherein said subcode comprises a quadriphase code.
8. A method of communicating among a plurality of stations comprising the steps of:
selecting a first subcode and a second subcode having low cross-correlation properties;
generating a first repeated codeword preamble code from a repeated series of said first subcode, said first repeated codeword preamble code comprising a first single continuous code;
generating a second repeated codeword preamble code from a repeated series of said second subcode said second repeated codeword preamble code comprising a second single continuous code; and
communicating among the plurality of stations using said first repeated codeword preamble code and said second repeated codeword preamble code as preambles preceding data messages;
generating a first odd-length code;
modifying a length of said first odd-length code by one chip to form said first subcode,
generating a second odd-length code; and
modifying a length of said second odd-length code by one chip to form said second subcode.
9. The method of claim 8 wherein said step of communicating among a plurality of stations using said first repeated codeword preamble code and said second repeated codeword preamble code comprises the step of modulating said first repeated
codeword preamble code and said second repeated codeword preamble code using an I and Q modulation format.
10. The method of claim 8 wherein said step of modifying a length of said first odd-length code by one chip comprises the step of truncating said first odd-length code by one chip to form said first subcode, and wherein said step of modifying a
length of said second odd-length code by one chip comprises the step of truncating said second odd-length code by one chip to form said second subcode.
11. The method of claim 8 wherein said step of modifying a length of said first odd-length code by one chip comprises the step of adding one chip to said first odd-length code to form said first subcode, and wherein said step of modifying a
length of said second odd-length code by one chip comprises the step of adding one chip to said second odd-length code to form said second subcode.
12. A communication system, comprising:
a first station, said first station comprising a transmitter for transmitting a preamble code and a data message, said preamble code comprising a kronecker product of a first subcode and a second subcode; and
a second station, said second station comprising a receiver for receiving said preamble code, synchronizing in response to said preamble code, and receiving said data message;
wherein said receiver of said second station comprises
a first correlator for correlating said preamble signal against said first subcode and generating a first correlation output signal in response thereto, said first correlation output signal comprising a first correlator correlation pulse
corresponding to each occurrence of said first subcode or its inverse in said preamble code; and
a second correlator for correlating said first correlation output signal against said second subcode and generating a second correlation output signal in response thereto, said second correlation output signal comprising a second correlator
correlation pulse when said preamble code is fully received.
13. A station for communication, comprising:
a receiver for receiving a repeated codeword preamble code and a data message, said repeated codeword preamble code comprising a repeated series of the same subcode;
a correlator for generating a plurality of correlation spikes in response to receiving said repeated codeword preamble code, each of said correlation spikes corresponding to a separate appearance of said subcode;
means for detecting receipt of said repeated codeword preamble code upon simultaneous detection of more than one of said correlation spikes; and
means for synchronizing the receiver in response to said repeated codeword preamble code.
14. The station for communication of claim 13, wherein said means for detecting receipt of said repeated codeword preamble code upon simultaneous detection of more than one of said correlation spikes comprises
an alert signal generator for generating an alert signal based on a maximum received correlation energy for said plurality of correlation spikes, and
a confirm signal generator for generating a confirm signal based on a minimum energy level of each of said plurality of correlation spikes.
15. The station for communication of claim 13, wherein said means for detecting receipt of said repeated codeword preamble code upon simultaneous detection of more than one of said correlation spikes detects a predefined number of said plurality
of correlation spikes.
16. The station for communication of claim 13, further comprising means for performing a verification of received data integrity of said data message following said repeated codeword preamble code.
17. The station for communication of claim 13, wherein said correlator comprises a mismatched filter for receiving said repeated codeword preamble code, said mismatched filter configured to detect said subcode.
18. The station for communication of claim 13, wherein said subcode comprises a Gold code.
19. The station for communication of claim 13, wherein said subcode comprises a member of a code family selected from the group of code families consisting of: Barker codes, Boztas codes, minimum peak sidelobe codes, and Kasami codes.
20. The station for communication of claim 13, wherein said subcode comprises a quadriphase code.
21. A communication system, comprising:
first station, said first station comprising a first transmitter for transmitting a first concatenated code generated from a kronecker product of a first subcode and a second subcode;
a second station, said second station comprising a second transmitter for transmitting a second concatenated code generated from a kronecker product of a third subcode and a fourth subcode, said second concatenated code having the same length as
said first concatenated code;
a first receiver, said first receiver configured to receive said first concatenated code; and
a second receiver, said second receiver configured to receive said second concatenated code.
22. The communication system of claim 21, wherein said first subcode and said third subcode are both Barker codes.
23. The communication system of claim 21, wherein said first subcode and said third subcode comprise the same code.
24. The communication system of claim 21, wherein each of said first subcode and said second subcode have a cross-correlation of no more than -10 dB with respect to each of said third subcode and said fourth subcode.
25. A communication system, comprising:
a first station, said first station comprising a first transmitter for transmitting a first concatenated code generated from a kronecker product of a first subcode and a second subcode, said first subcode being an inner code in said kronecker
product and said second subcode being an outer code in said kronecker product;
a second station, said second station comprising a second transmitter for transmitting a second concatenated code generated from a kronecker product of said first subcode and said second subcode, said first subcode being the outer code in said
second kronecker product and said second subcode being the inner code in said second kronecker product, said second concatenated code having the same length as the first concatenated code;
a first receiver, said first receiver configured to detect said first concatenated code; and
a second receiver, said second receiver configured to detect said second concatenated code.
26. A communication system, comprising:
a first station, said first station comprising a first transmitter for transmitting a first concatenated code generated from a product of a first subcode and a first Barker code;
a second station, said second station comprising a second transmitter for transmitting a second concatenated code having the same length as said first concatenated code, said second concatenated code generated from a product of a second subcode
and a second Barker code;
a first receiver, said first receiver configured to detect said first concatenated code; and
a second receiver, said second receiver configured to detect said second concatenated code.
27. The communication system of claim 26 wherein said first Barker code and said second Barker code comprise the same code.
28. The communication system of claim 26 wherein said first subcode and said second subcode have a cross-correlation of no more than -10 dB.
29. A communication system, comprising:
first station, said first station comprising a first transmitter for transmitting a first repeated codeword preamble code generated from a repeated series of a first subcode;
a second station, said second station comprising a second transmitter for transmitting a second repeated codeword preamble code generated from a repeated series of a second subcode, said second subcode having relatively low cross-correlation
properties with respect to said first subcode;
a first receiver, said first receiver configured to detect said first repeated codeword preamble code; and
a second receiver, said second receiver configured to detect said second repeated codeword preamble code.
30. The communication system of claim 29, wherein said first subcode and said second subcode each comprises a quadriphase code.
31. The communication system of claim 30, wherein each quadriphase code comprises a Boztas code.
32. The communication system of claim 29, wherein said first subcode comprises a first odd-length code modified in length by one chip, and wherein said second subcode comprises a second odd-length code modified in length by one chip.
33. The communication system of claim 32, wherein said first transmitter and said second transmitter each comprise means for modulating said first repeated codeword preamble code and said second repeated codeword preamble code, respectively,
using an I and Q modulation format.
34. A communication system, comprising:
first station, said first station comprising a first transmitter for transmitting a first preamble code and a first data message, said first preamble code generated from a series of a first subcode;
a second station, said second station comprising a second transmitter for transmitting a second preamble code and a second data message, said second preamble code generated from a series of a second subcode, wherein said second subcode has the
same length as said first subcode and a low cross-correlation level with respect said first subcode, and wherein said first preamble code and said second preamble code have the same length;
a first receiver for receiving said first preamble code and said first data message, and for synchronizing to said first preamble code; and
a second receiver for receiving said second preamble code and said second data message, and for synchronizing to said second preamble. |
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Description |
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MICROFICHE APPENDIX
A Microfiche Appendix is filed herewith which comprises one microfiche sheet of 31 frames.
FIELD OF THE INVENTION
The field of the present invention relates to communications and, more particularly, to an improved preamble code structure and method for code detection for use in a wifeless communication system.
BACKGROUND
Wireless communication systems typically comprise a number of mobile "user stations" or "handsets" and a number of stationary or fixed "base stations" which are capable communicating with each other. The base stations and user stations may
communicate using frequency division multiple access (FDMA), wherein transmissions are distinguished by using different assigned frequencies; time division multiple access (TDMA), wherein transmissions are distinguished according to assigned time slots
within a time frame; code division multiple access (CDMA), wherein transmissions are distinguished according to different assigned codes; or various combinations thereof.
One type of communication used in wireless applications is spread spectrum communication, wherein the bandwidth of the signal being transmitted generally exceeds the bandwidth required to transmit the data being sent. In spread spectrum
communication, a data signal is typically modulated with a pseudo-random chip code to generate a transmitted signal spread over a relatively wide bandwidth. At the receiver the received spread spectrum signal is despread in order to recover the original
data. One method of despreading of the spread spectrum signal is by correlating the received signal with a reference code matching the pseudo-noise code used by the transmitter to transmit the data. After initial correlation is achieved, in many
systems it is necessary to track the incoming signal so as to maintain synchronization and keep it time-aligned with the local reference code to allow continued despreading of the received signal.
In order to carry out communication between a base station and a user station, a communication link must first be established. In a TDMA system, a communication link may comprise, e.g., a time slot having a forward link portion and a reverse
link portion wherein a base station and a user station exchange communications in time division duplex. Establishment of the communication link can be difficult in a spread-spectrum TDMA communication system, due to the length of time that may be
required to synchronize the transmitter and the receiver as well as the relative brevity of the time slot within which synchronization can take place.
In order to assist rapid synchronization of communication in spread spectrum and other communication systems, a preamble code preceding an information message may be used. A preamble code may comprise a relatively easily identifiable code
sequence that marks the start of the information message and thereby allows the transmitter and receiver to synchronize. The receiver searches for the preamble code and, after locating it, knows when to expect the remaining information message and what
timing adjustments may need to be made for optimum correlation of the information message.
Use of a preamble code can be particularly advantageous in a TDMA system because of the intermittent nature of the periodic transmissions between a base station and the user stations, which may require re-synchronization each time frame or series
of time frames. Because the length of each burst is inherently limited by the duration of a time slot (or time slots), information is transmitted to and from a given user station in a TDMA system periodically over a series of time frames, with the base
station and user station typically communicating only once per time frame, during a specified time slot. Due to the periodic nature of TDMA transmissions over a given link, the base station and the user station using the link have to look for the
intermittent messages sent to them, which may be separated in time by an entire time frame or even more (e.g., several time frames) in some cases.
The fact that the user stations can be mobile may cause the periodic transmissions to drift within the allocated time slot. In addition, there is the possibility of drift between transmitter and receiver clocks. Thus, the receiver may not know
precisely at what point the incoming burst will be received, although in some cases the expected time of arrival may be narrowed down to within a predefined window around the start of the time slot. When the receiver is waiting for its designated
message, it may receive extraneous messages from nearby users of the same frequency spectrum or neighboring frequency spectrum, or may otherwise receive interference or noise and mistakenly interpret it as part of the message designated for it. A
preamble code helps minimize possible misidentification of noise or interference as a valid message by assisting in the detection of the start of a designated burst. To prevent confusion at the user station, the preamble code for a given user station
must be distinguished from the preamble codes as well as any other codes that may be targeted to any other the user station (or to the base station) during a time slot when the given user station is to communicate with the base station.
A preamble code may need to be identified rapidly, such as where a time slot is relatively short. This requirement generally suggests the use of short preamble codes. At the same time, a preamble code is preferably resistant to noise,
interference and multipath effects, as well as false alarms due to autocorrelation peaks and cross-correlations, so as to ensure the highest probability of proper detection and identification of the preamble code at the receiver. If a preamble code is
not properly identified by the receiver, the entire information message for the burst being sent may be lost.
One option to increase likelihood of preamble code detection is to increase the power of the transmitted preamble code over that of the transmit power for the information message, thereby increasing signal-to-noise ratio of the transmitted
preamble code. While increasing the transmit power for the preamble code may decrease sensitivity to noise and interference, higher power transmissions for preamble codes may unduly interfere with users of the same or neighboring frequency spectra.
Moreover, in certain low power applications (such as various types of handsets), it may not be feasible to increase the transmission power of preamble codes. Even if feasible, increasing the transmit power could cause early depletion of battery charge
for some mobile handsets.
Another option is to increase the length of the preamble code so as to provide better discrimination as against noise and other signals. However, merely elongating the preamble code generally leads to more complex synchronization filters and
increases the time needed to detect the preamble code.
Accordingly, it would be advantageous to provide a preamble code well suited to a TDMA communication system or other communication system requiring rapid synchronization at the receiver. It would further be advantageous to provide such a
preamble code while maintaining a relatively simple synchronization filter structure. It would further be advantageous to provide a preamble code having resistance to noise and interference, without necessarily requiring increased transmission power.
As a further consideration in preamble code design, a preamble code may be used for selection of an antenna channel in a system where antenna diversity is employed. The received signal quality of the preamble code is evaluated for each antenna
branch, and the best antenna or set of antennas is selected to receive the information message. Thus, a preamble code is preferably constructed so as to be well suited for use in a system employing antenna diversity, and to allow relatively easy
evaluation of received signal quality so as to facilitate antenna selection.
SUMMARY OF THE INVENTION
The invention in one aspect comprises a concatenated preamble code structure and means for detecting a concatenated code such as a concatenated preamble code.
In a first embodiment of the invention, a concatenated preamble code is formed by a kronecker product between two subcodes of different lengths. At the receiver a two-stage processor is preferably used to detect the concatenated preamble code.
The two-stage processor preferably comprises a series of two correlation filters. A family of concatenated preamble codes may be generated from the kronecker product of a first subcode (such as a Barker code) with each of a plurality of other subcodes,
each of these other subcodes preferably having low cross-correlation levels with respect to one another.
In a second embodiment of the invention, a concatenated preamble code is formed by a kronecker product between two subcodes of approximately equal length. The two selected subcodes used to form the concatenated preamble code are preferably short
and preferably have relatively low a periodic auto-correlation responses.
In another aspect of the invention, a plurality of correlation filters including at least one mismatched filter are employed in series for detecting a concatenated code, such as a concatenated preamble code. The mismatched filter is preferably
the first in the series of filters and is preferably followed by a single matched filter. Odd length Barker codes may be used to reduce processing gain losses attributed to use of a mismatched filter. Alternatively, to support a modulation format using
in-phase (I) and quadrature (Q) baseband components, an even code length may be selected by adding a chip to or truncating a chip from an existing odd code (such as a Gold code or a Kasami code).
In another aspect of the invention, a repeated codeword preamble. (RCP) code is formed by transmitting a single short codeword (i.e., subcode) several times in a row. At the receiver a relatively simple matched filter is used to generate a
series of spikes separated by the period of the subcode. An alert/confirm detector may be used to non-coherently add together individual correlation spikes and reject false alarms. The alert function preferably measures total correlation energy and
adjusts detection thresholds based thereon, while the confirm function preferably ensures that the proper number of correlation spikes exceed a detection threshold at the correct times.
The various preamble code structures and means for detection thereof are disclosed with respect to a preferred TDMA communication system, wherein a plurality of user stations communicate with a base station, one in each time slot (or virtual time
slot) in time division duplex.
Other further variations and refinements are also disclosed herein.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of a communication system comprising user stations and base stations.
FIG. 2 is a block diagram of a communication network including user stations and base stations.
FIG. 3 is a diagram of a preferred cellular environment according to one or more aspects of the present invention.
FIG. 4 is a timing diagram of a polling loop illustrating an exemplary over-the-air protocol.
FIG. 5 is a diagram illustrating a protocol for communication.
FIG. 6A is a timing diagram of an exemplary time slot structure.
FIGS. 6B and 6C are timing diagrams of a base transmit data frame structure and a mobile station transmit data frame structure, respectively.
FIG. 7 is a timing diagram of a polling loop illustrating an alternative over-the-air protocol. FIG. 8 is a circuit diagram of a preferred non-coherent CPM correlator.
FIG. 9 is a diagram showing the construction of concatenated preamble codes.
FIG. 10 is a block diagram of a two-stage correlator using two matched filters for detecting a concatenated preamble code.
FIGS. 11A and 11B are graphs of responses of a two-stage matched filter correlator to a concatenated preamble code derived from a Barker code and a minimum peak sidelobe code.
FIG. 12 is a graph depicting a two-stage matched filter correlator output for a Barker-11 code concatenated with itself.
FIGS. 13A-D and 14A-D are graphs illustrating correlation responses of a two-stage correlator using matched filters to detect concatenated preamble codes derived from a Barker code and a length-32 minimum peak sidelobe code.
FIGS. 15A-D are graphs illustrating correlation properties of two length-11 Barker codes.
FIGS. 16A-D are graphs illustrating correlation responses of a two-stage correlator using matched filters to detect concatenated preamble codes derived from the length-11 Barker codes used to generate the graphs of FIG. 15A-D.
FIGS. 17A-D are graphs illustrating correlation responses to two other concatenated preamble codes derived from the length-11 Barker codes used to generate the graphs of FIGS. 15A-D.
FIG. 18 is a block diagram of a two-stage correlator using a mismatched filter and matched filter for detecting a concatenated preamble code.
FIGS. 19A and 19B are graphs comparing matched filter and mismatched filter responses to a Barker-4 code.
FIGS. 20A and 20B are graphs showing matched filter response where quantizing is employed, for different levels of quantizing.
FIGS. 21A-D and 22A-D are graphs illustrating correlation responses of a two-stage mismatched filter correlator to concatenated preamble codes derived from a Barker code and a length-32 minimum peak sidelobe code.
FIGS. 23A-D are graphs showing the responses of various matched and mismatched filters having different filter structures to a particular length-4 code.
FIGS. 24A-D and 25A-D are graphs illustrating correlation responses of a two-stage mismatched filter correlator including the mismatched filter of FIG. 23D to concatenated preamble codes derived from a Barker code and a length-32 minimum peak
sidelobe code.
FIGS. 26 and 27 are graphs comparing the responses of matched and mismatched filters to a length-140 concatenated preamble code generated from the kronecker product of a Barker-5 (B5) code and an MPS28 code.
FIGS. 28 through 31 compare the response of a matched filter with the responses of various mismatched filters to a length-5 Barker code.
FIGS. 32A-D and 33A-D are graphs illustrating correlation responses of a two-stage mismatched filter correlator including the mismatched filter of FIG. 31 to concatenated preamble codes derived from a length-5 Barker code and a length-25 minimum
peak sidelobe code.
FIGS. 34 through 37 compare the responses of a matched filter and various mismatched filters to a length-3 Barker code.
FIGS. 38A-D and 39A-D are graphs illustrating correlation responses of a two-stage mismatched filter correlator including the mismatched filter of FIG. 37 to concatenated preamble codes derived from a length-3 Barker code and a length-39 code.
FIGS. 40A-D are graphs illustrating correlation responses of matched filters configured for length-124 repeated codeword preamble codes constructed from length-31 Gold codes.
FIGS. 41A-D are graphs illustrating the response of a Plagge filter to the Gold-31 preamble codes from FIGS. 40A-D.
FIGS. 42A-D and 43A-D are graphs illustrating correlation responses of matched filters to length-126 repeated codeword preamble codes constructed from Kasami-63 codes.
FIGS. 44A--D and 45A-D are graphs illustrating correlation responses of matched filters to length-128 preamble codes constructed from augmented Gold codes.
FIGS. 46A-D and 47A-D are graphs illustrating correlation responses of matched filters to length-120 repeated codeword preamble codes constructed from truncated Gold codes.
FIGS. 48A-D are graphs illustrating correlation responses of matched filters to length-128 repeated codeword preamble codes constructed from augmented Kasami codes.
FIGS. 49A-D are graphs illustrating correlation responses of matched filters to length-124 repeated codeword preamble codes constructed from truncated Kasami codes.
FIGS. 50A-D and 51A-D are graphs illustrating correlation responses of matched filters to length-60 repeated codeword preamble codes constructed from Boztas codes.
FIGS. 52A-D, 53A-D and 54A-D are graphs illustrating correlation responses of matched filters to various length quadriphase preamble codes constructed from Boztas codes.
FIGS. 55A-D and 56A-D are graphs comparing simulated correlation responses of a length-128 non-repeating preamble code with a length-120 repeated codeword preamble code in the presence of noise and interference.
FIG. 57 is a block diagram of one embodiment of a repeated codeword preamble code detector.
FIGS. 58A and 58B are charts comparing performance of preamble code detection with and without partitioning, for various levels of detection probability and with different false alarm probabilities.
FIG. 59 is a block diagram of a preferred alert/confirm preamble code detector, and
FIG. 59A is a block diagram of a preferred moving average filter for use in the FIG. 59 preamble code detector.
FIGS. 60A-D, 61A-D and 62A-D are graphs showing exemplary signal output levels in accordance with the FIG. 59 alert/confirm preamble code detector.
FIGS. 63A and 63B are charts comparing expected preamble code detection sensitivities with and without partitioning, for various levels of detection probability and with different false alarm probabilities, based on evaluation of a Generalized
Marcum's Q function.
FIG. 64 is a graph comparing an additional mean signal strength necessary to maintain a performance level based on various numbers of antenna channel paths and rake channel paths.
FIG. 65 is a chart comparing preamble threshold detection setting requirements for various preamble code lengths and alternative probability of detection figures.
FIG. 66 is a block diagram of a mismatched filter with quantized coefficients.
FIG. 67 is a detailed block diagram of a spread spectrum correlator.
FIGS. 68A and 68B are tables summarizing various properties relating to Gold codes and Kasami codes.
FIGS. 69A through 69D are tables summarizing various properties relating to particular quadriphase codes.
FIG. 70 is a table of eight length-63 Kasami codes.
FIG. 71 is a table of four length-15 quadriphase Boztas codes in Galois representation and their seed values.
FIG. 72 is a table of two length-30 Grey-mapped Boztas codes and their seed values.
FIG. 73 is a table of three length-31 quadriphase Boztas codes in Galois representation and their seed values.
FIG. 74 is a block diagram of a transistor using an I and Q modulation technique.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Various aspects of the invention, including different preamble code structures and means for detecting preamble codes, are described with reference to a preferred TDMA communication system as set forth below.
FIG. 1 illustrates the cellular arrangement of a communication system 101 comprising one or more user stations 102 and one or more base stations 104 arranged within a plurality of communication cells 103. Each base station 104 and user station
102 preferably comprises one or more radios each comprising a receiver, a transmitter, and one or more antennas. Each cell 103 preferably includes a single base station 104 located near the center of the cell 103.
FIG. 2 is a diagram showing additional details of a communication system architecture. The system architecture includes a plurality of base stations 104 which communicate with a plurality of user stations 102. Each base station 104 is shown
coupled to a base station controller 105 by any of a variety of communication paths 109. The communication paths 109 each comprise one or more communication links 110 (e.g., a coaxial cable, a fiber optic cable, a digital radio link, or a telephone
line). Each base station controller 105 is preferably connected to one or more communication networks 106 such as a public switched telephone network (PSTN) or personal communication system switching center (PCSC). Each base station controller 105 is
shown connected to the communication network(s) 106 by means of one or more communication paths 108, each of which may include one or more communication links 110, examples of which are explained above. The communication system architecture shown in
FIG. 2 also may include one or more "intelligent" base stations 107 which connects a user station 102 directly to a communication network 106 without interfacing through a base station controller 105.
In a preferred embodiment, communication between base stations 104 and user stations 102 is accomplished using spread spectrum technology. A preferred method of demodulating and correlating a continuous phase modulated (CPM) spread spectrum
signal is described in U.S. Pat. No. 5,659,574 filed Jun. 7, 1995, hereby incorporated by reference as if set forth fully herein. FIG. 8 is a circuit diagram of a preferred non-coherent continuous phase modulated (CPM) correlator that may be used for
detecting a preamble code or other spread spectrum code. The operation of the FIG. 8 circuit is more fully described in the U.S. Pat. No. 5,659,574 referenced immediately above.
Other exemplary correlators are described in, e.g., U.S. Pat. Nos. 5,022,047 and 5,016,255, each of which are assigned to the assignee of the present invention, and each of which are incorporated herein by reference as if fully set forth
herein.
Preferably, the base station 104 and user station 102 of FIG. 1 establish synchronization and communication using an M-ary direct sequence spreading technique in which multiple bits of data are transmitted for each spread spectrum symbol.
Suitable M-ary spread spectrum transmission and reception techniques are described in, e.g., U.S. Pat. No. 5,022,047 or in U.S. Pat. No. 5,757,847 1995, both of which are incorporated herein by reference as if set forth fully herein. In a preferred
embodiment, the base station 104 and user stations 102 each transmit an M-ary direct sequence spread spectrum signal, with M=32, using spread spectrum codes-(called "symbol codes") of the same length (e.g., 32 chips each in length). In a preferred M-ary
spread spectrum modulation technique, thirty-two different symbol codes are used to represent up to thirty-two different data symbols, each comprising five bits of data, with differential phase encoding allowing transmission of a 6th bit of data for each
symbol code. Techniques of phase encoding for transmission of an additional bit of information per symbol code are described in, e.g., U.S. Pat. No. 5,757,847 referenced above.
FIG. 3 is a diagram of a preferred cellular environment in which the invention may operate.
In FIG. 3, a geographical region 301 is divided into a plurality of cells 103. Associated with each cell 103 is an assigned frequency Fx and an assigned spread spectrum code Cy. Preferably, three different frequencies F1, F2 and F3 are assigned
in such a manner that no two adjacent cells have the same assigned frequency F1, F2 or F3. The effect of such a frequency reuse pattern is the minimization of interference between adjacent cells with a minimum number of frequencies used. The preferable
frequency reuse pattern is F=3.
To further reduce the possibility of intercell interference, different near-orthogonal spread spectrum codes C1 through C7 are assigned as shown in a repeating pattern overlapping the frequency reuse pattern. Although a repeating pattern of
seven spread spectrum codes C1 through C7 is preferred, a pattern involving other numbers of spread spectrum codes may be suitable depending upon the particular application. Further information regarding a suitable cellular environment for operation of
the invention may be found in U.S. Pat. No. 5,402,413 which is incorporated herein by reference as if fully set forth herein.
The use of spread spectrum techniques for carrier modulation permits a frequency reuse factor of N=3 for allocating different carrier frequencies F1, F2 and F3 to adjacent cells 103. Interference between cells 103 using the same carrier
frequency F1, F2 or F3 is reduced by the propagation loss due to the distance separating the cells 103 (no two cells 103 using the same frequency F1, F2 or F3 are less than two cells 103 in distance away from one another), and also by the spread spectrum
processing gain of cells 103 using the same carrier frequencies F1, F2 or F3, obtained by the use of near-orthogonal spreading codes.
The invention may be used, as explained hereinafter, in conjunction with antenna diversity techniques. For example, a preamble code such as described herein may be used to sound a channel and allow selection of one from a plurality of antennas
for the message following the preamble code.
Different types and numbers of antennas may be connected to the base station 104, depending on the type of application. For low density suburban or rural applications an omnidirectional antenna is preferable to provide maximum coverage with the
fewest base stations 104. For example, a vertically polarized omnidirectional antenna may be employed having a gain of approximately 9 dB. The 9 dB of gain permits a relatively large radius cell even with an omnidirectional azimuthal pattern.
A single steered phased array antenna is presently preferred for applications requiring a high gain, highly directional antenna. For example, to permit a single base station 104 to cover large, sparsely populated area, a steered phased array
antenna with up to 20 dB of horizontal directivity is presently preferred. The steered phased array antenna preferably utilizes circular polarization so that high level delayed clutter signals reflected from buildings or other obstructions within the
beam path do not interfere with the received signals from the user stations 102. Because reflected signals are typically reversed in polarization, they will be largely rejected by circularly polarized antennas.
In suburban and low density urban areas, directional antennas with 120 degree azimuth beamwidths and 13 dB vertical gain are presently preferred so that a cell 103 can be divided into three sectors, with each sector accommodating a full load of
user stations (e.g., 16 or 32 user stations 102). The use of high gain, directional antennas reduces the delay spread in severe multipath environments by rejecting multipath components arriving from outside the main beam of the antenna. Additionally,
directional antennas reduce interference to user stations 102 in neighboring cells and fixed microwave facilities which may be operating nearby.
In more dense urban areas and other areas with significant multipath problems, the number of directional antennas used by a base station 104 is preferably increased to provide antenna diversity as a means of combatting signal degradations from
multipath propagation. When multiple antennas are employed, circuitry for selecting an antenna for each transmission which best matches the communication channel between the base station 104 and user station 102 is preferred.
In one embodiment, the user station 102 employs a halfwave dipole antenna which is linearly polarized and provides a gain of 2 dB with an omnidirectional pattern perpendicular to the antenna axis. At a nominal frequency of 1900 MHz, a half
wavelength is approximately 3 inches, which is fitted within a handset.
Details of a particular protocol suitable for low power pocket phone operation in a PCS microcell environment is found in U.S. Pat. No. 5,597,080, hereby incorporated by reference as if set forth fully herein.
In a preferred embodiment, the user stations 102 and base stations 104 communicate using time division multiple access (TDMA) techniques and preferably time division duplexing (TDD). According to these techniques, a continuous series of polling
loops or major time frames for organizing communication between a base station 104 and user stations 102 is provided.
FIG. 4 is a diagram of a polling loop 401 (also referred to as a major time frame). In FIG. 4 each polling loop 401 is further divided into multiple time slots 410 which are assigned for communication between base stations 104 and user stations
102. A base station 104 may communicate with a plurality of user stations 102 on a periodic basis over consecutive polling loops 401. In a preferred embodiment, the polling loop 401 is divided into sixteen time slots 410, and each time slot 410 has a
duration of 1.25 milliseconds.
Preferred features of a time slot 410 are depicted in FIG. 5. As shown in FIG. 5, a time slot 410 of polling loop 401 comprises a user station transmit frame 515, a guard time interval 535, and a base station transmit frame 545. A base station
104 transmits a base station transmission 440 during the base station transmit frame 545 to a user station 102 with which the base station 104 is communicating. The user station 102 transmits to the base station 104 a user station transmission 430
during the user station transmit frame 515.
Time division duplex permits common antennas to be used for transmit and receive functions at both the base station 104 and the user stations 102, generally without the need for-antenna diplexers. Common antennas can be used to transmit and
receive because these functions are separated in time at each of the user stations 102 and base stations 104. The use of common antennas generally results in simplicity of the design of the base station 104 and user station 102.
FIGS. 6A through 6C illustrate preferred time slot components, and the relative location of the preamble code (or preamble codes) with respect to the other time slot components.
FIG. 6A shows the structure of a preferred structure of a time slot 601 comprising a variable radio delay gap 605, a user station transmit field 610, a base processor gap 615, a guard time 620, a base transmit field 625, and a radar gap 630.
Each user station transmit field 610 comprises a user preamble code field 635, a user preamble sounding gap 640, and a user station data field 645. Similarly, each base transmit frame 625 comprises a base preamble code field 650, a base preamble
sounding gap 655, and a base transmit data field 660. The user preamble code field 635 and base preamble code field 650 are illustrated in more detail in a variety of different embodiments described later herein.
FIG. 6B shows a preferred structure for the base transmit data field 660. The base transmit data field 660 comprises a base header field 665, a base D-channel field 670, a base data field 675, and a base cyclical redundancy check (CRC) field
680. In a preferred embodiment, the base header field 665 is 23 bits, the base D-channel field 670 is 8 bits, the base data field 625 is 192 bits, and the base CRC field 680 is 16 bits.
FIG. 6C shows a preferred structure for the user station transmit data field 645. The user station data field 645 comprises a user header field 685, a user D-channel field 690, a user data field 695, and a user CRC field 697. In a preferred
embodiment, the user header field 685 is 17 bits, the user D-channel field 690 is 8 bits, the user data field 695 is 192 bits, and the user CRC field 697 is 16 bits.
FIG. 7 shows an alternative polling loop structure that may be used in accordance with one or more embodiments of the present invention, and is more fully described in U.S. Pat. No. 6,005,856, And incorporated herein by reference as if fully
set forth herein. Further details regarding the FIG. 4 polling loop structure may also be found in that application. Details regarding a presently preferred method for establishing communication between a base station 104 and a user station 102 may be
found in U.S. Pat. No. 5,455,822, incorporated herein by reference as if set forth fully herein, or in the copending application referenced immediately above.
Several embodiments of the invention make use of one or more matched filter(s) and/or one or more mismatched filter(s). Matched filters for spread-spectrum communication (including M-ary spread spectrum communication) are known in the art as
exemplified by, e.g., U.S. Pat. Nos. 5,022,047 and 5,016,255, both of which are assigned to the assignee of the present invention and incorporated by reference as if fully set forth herein. An exemplary matched filter 1721 (i.e., correlator) which
may be used to detect binary phase shift keyed (BPSK) modulated spread spectrum signals is shown in FIG. 67. Its operation is described in detail in U.S. Pat. No. 5,022,047. A preferred correlator for detecting a continuous phase modulated (CPM)
spread spectrum signal is described in U.S. Pat. No. 5,659,574, previously incorporated herein by reference.
A variety of other types of correlators and/or matched filters are known in the art and would be suitable in various embodiments of the invention set forth herein. Selection of a suitable correlator depends on a variety of factors, including the
type of modulation used to transmit the signal to be detected.
FIG. 66 is a diagram of a representative mismatched filter 1501 configured to detect a particular length-4 code sequence of [1 1 -1 1]. The mismatched filter 1501 is essentially a variety of finite impulse response (FIR) filter, and has a
plurality of delay stages 1505 connected in series, along which an input signal 1502 propagates. Each delay stage 1505 is preferably one chip interval in length (although non-integral chip spacings may also be used). Outputs from a plurality of delay
stages 1505 are connected to a plurality of multipliers 1507 which multiply the delay stage output by the value of the coefficient associated therewith. Outputs from each of the multipliers 1507 are connected to a summer 1509, which sums its inputs to
arrive at a correlation output signal 1510.
The construction of mismatched filters, including derivation of appropriate filter coefficients, is well known in the art of signal detection. Mismatched filters can be constructed for most code types, and are not limited to Barker codes.
Appropriate coefficients for a mismatched filter to detect an aperiodic code may be determined by using, for example, a simplex method or gradient method. Optimizing computer programs-are also available to assist in derivation of suitable filter
coefficients. The mismatched filter may be constructed to operate in either an analog fashion (e.g., using a SAW device) or a digital fashion. Generally, the mismatched filter coefficients may have as values any real or complex number.
The coefficients for mismatched filters are, in some embodiments described herein, quantized to values from a limited set of coefficients. For N-bit quantization, 2.sup.N possible quantized values are achievable. For example, using 2-bit
quantization, the magnitude of each coefficient takes on one of 2.sup.2 (i.e., four) values. Using 3-bit quantization, the magnitude of each coefficient takes on one of 2.sup.3 (i.e., eight) values. The largest coefficient magnitude is given the
largest quantization value; the rest of coefficient magnitudes are reduced in the same proportion as the largest (i.e., normalized) and rounded to the nearest quantized value. The sign of the coefficient is retained separately.
The FIG. 66 mismatched filter uses 13 stages to det | | |
