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BACKGROUND OF THE INVENTION
MICROFICHE APPENDIX
This application includes a microfiche appendix consisting of 32 slides and 6,267 frames, which is a copy of the provisional application under which priority is claimed and updated source code.
1. Field of the Invention
This invention relates to phacoemulsification devices and, more particularly, to a method for controlling a phacoemulsification device.
2. Related Art
Ultrasonic probes have traditionally been used for phacoemulsification, namely, for rupturing of cataracts in the eye and for aspiration of the pieces of tissue disrupted. These ultrasonic probes must be carefully powered for proper operation.
Operating the ultrasonic probe at its resonant frequency takes advantage of the resonant characteristics of the ultrasonic transducer. Resonance is defined as the phenomenon wherein a system is driven at or near one of its natural modes.
Accordingly, the prior art has focused on how to determine the resonant frequency of a transducer. Theoretically, this problem has been solved. A typical way of determining the resonant frequency of an ultrasonic transducer is to compare the
phase angle between the voltage waveform applied to the ultrasonic transducer and the waveform of the current drawn by the transducer.
When voltage is applied to a circuit, current will flow through the circuit. When the voltage and current waveforms are viewed for a particular circuit, the current waveform will lag the voltage waveform if the circuit is inductive, and the
voltage waveform will lag the current waveform if the circuit is capacitive. The time difference between the points when the current waveform and voltage waveform intersect the zero axis is measured in trigonometric terms by the phase angle .PHI.. For
purely resistive circuits, .PHI. equals zero and the voltage in the current waveforms are said to be in phase. For purely inductive circuits, .PHI. equals 90.degree. and for purely capacitive circuits, .PHI. equals -90.degree. and the voltage in
the current waveforms are said to be out of phase.
The presence of an inductive or capacitive reactance component in a load impedance will decrease the efficiency of power delivery of the system since only resistive components can actually dissipate power.
For circuits containing all three elements, resistors, inductors and capacitors, there will be some frequencies where the total impedance of the circuit will appear purely resistive even though the circuit contains reactive elements, i.e., the
resistive elements plus the imaginary component caused by the presence of the inductive and capacitive elements. These frequencies are at or near the resonant and/or anti-resonant frequencies.
Therefore, in theory, one method of determining the resonant frequencies of certain types of complex circuits is to apply an alternating voltage to the circuit and to vary the frequency until the phase angle .PHI. between the voltage and current
is zero. The frequencies where this condition occurs are the actual resonant frequencies of that particular circuit. The resonant frequency is that frequency or frequencies at which the circuit response (i.e., admittance) is locally a maximum, and the
anti-resonant frequency is that frequency or frequencies at which the response achieves a local minimum.
When driving a circuit having both resistive and reactive components, it is important to know the value of the phase angle .PHI. because the power delivered to a load is given by the following equation:
where V is the voltage drop across the load impedance; I is the series current flowing through the load impedance; and cosine phi is the power factor of the circuit. Clearly, for a phase angle equal to zero, cosine (0) equals 1 and the transfer
of power from the source to the circuit is at maximum. This situation exists where a purely resistive load exists.
As these theoretical principles are practically applied, problems have been encountered. Specifically, as environmental conditions such as temperature, time, etc., change, the characteristics of the probe changes. These changes are reflected as
changes in the values of the various resistive and reactive components of the ultrasonic probe electrical model of FIG. 1. In other words, as the environmental factors change, the mechanical resonant frequency of ultrasonic probe changes also. To solve
this problem, there has been a direction in the prior art to provide a phase locked circuit to ensure that the phase angle of the system, .PHI., will be zero, such as for example in U.S. Pat. Nos. 5,446,416; 5,210,509; 5,097,219; 5,072,195; 4,973,876;
4,484,154; and 4,114,110.
However, loading on the transducer will have a damping effect on the vibrations of the transducer. In other words, the load may dampen the vibrations of the transducer. When this condition occurs, the resonant frequency may change and phase
angle .PHI. will longer be zero and the transfer of power will no longer be optimum. Therefore, unless provisions are made in the circuit to alter the phase angle .PHI., optimum power transfer cannot be achieved.
Accordingly, methods other than locking the phase angle .PHI. have been explored such as using a tunable inductor in a control system to cancel out the capacitive reactants of the load impedance presented by the ultrasonic transducer, such as
that disclosed in U.S. Pat. Nos. 4,970,656; and 4,954,960. Alternatively, using the admittance of the ultrasonic transducer as the tuning parameter rather than the phase angle has also been explored in U.S. Pat. No. 5,431,664.
Approaching this problem from a purely output power standpoint has also been explored in U.S. Pat. No. 5,331,951 in which the actual electrical power supplied to the drive circuit is examined and the supply voltage is varied after comparing the
electrical power supplied with the desired transducer power level. Tangentially, this patent also addresses a way to substantially minimize the power amplifier's power consumption by providing a boost regulator for supplying voltage to the amplifier.
In yet another approach, phase-regulated power and frequency control is utilized, such as in U.S. Pat. No. 4,849,872. Therein the initial resonance frequency of the ultrasonic transducer is determined and a capacitive phase angle between the
voltage waveform and current waveform is introduced and maintained so that by phase control of the phase control circuit, the operating frequency of the oscillator is reduced relative to the series resonance frequency of the transducer. The phase angle
is typically maintained as a non-zero constant. Similarly, in U.S. Pat. No. 4,888,565, a power control feedback loop for monitoring the output signal and a frequency control feedback loop are utilized to provide maximum current. This approach relies
on holding the mains current constant.
An electrical model of a ultrasonic phacoemulsification probe in the vicinity of resonance is provided in FIG. 1. The model has a voltage source 1401 connected to a 1130 picofarad capacitor 1402 connected in parallel to a series RLC circuit
1403, wherein the resistor is 220 ohms, the inductor is 1.708 henrys, and the capacitor is 18 picofarads.
When examining the apparent power resulting from the electrical model, the graphs of FIGS. 2 and 3 are obtained. As seen in these Figures, the apparent power peaks at 28.661 kHz with a phase angle of approximately -42 degrees. This is expected
due to the parallel capacitance in RLC circuit 1403.
When examining the real power resulting from the electrical model, the graphs of FIGS. 4 and 5 are obtained. As seen in these Figures, the real power peaks correctly at 28.7 kHz, but the phase angle is approximately -24.5 degrees.
When a compensating inductor with a calculated value of 27.21 millihenrys is placed in ghost block 1404 of FIG. 1 to cancel the reactive component of FIG. 1 and the resultant apparent power and real power information is obtained as in FIGS. 6 and
7, the apparent power and the real power now both correctly peak at 28.7 kHz with a phase of approximately -0.5 degrees. Thus, it can be seen that the inductor in ghost block 1404 compensates the parallel capacitance 1402 and makes the circuit appear
resistive (zero phase) at resonance. From these graphs, it is clear that the real power provides a more accurate view of resonance frequency, unless a compensating inductor is added near resonance. Accordingly, resonant frequency is defined herein as
the frequency at which real power achieves a (local) maximum. However, apparent power may be used to determine the resonant frequency if the parallel capacitance is compensated at resonance. Apparent power provides an approximation of resonant
frequency (frequency at which the local maximum occurs) if a compensating inductor compensates parallel capacitance 1402 near resonance.
Therefore, there is a need in the art to maximize the power output to an ultrasonic transducer which is responsive to both environmental changes as well as changes in loading, and yet which also does not necessarily require a fixed phase angle or
a constant current.
SUMMARY OF THE INVENTION
It is in the view of the above problems that the present invention was developed. The invention is an improved phacoemulsification probe drive circuit for supplying electrical power to an ultrasonic transducer. The drive circuit has a power
control loop and a frequency control loop. The power control loop has a variable gain amplifier whose output is an input to a power amplifier. After the power amplifier amplifies power, power is delivered to a transformer and, thereafter, to a
transducer. The voltage and current applied to the primary of the transformer are sensed to generate a signal proportional to the power (real or apparent) and the result is compared against a power command originating from a foot pedal. Once compared,
the result of this comparison is sent to a first controller which acts upon the information by sending a corrective signal to the variable gain amplifier. Also, the phase of the voltage and current waveforms applied to the primary of the transformer are
sensed by a phase detector. The phase angle is then derived and compared against a phase command which is determined from the initial calibration of the system. The summer/difference block sends its resulting comparison to a second controller which
sends a control signal to the voltage controlled oscillator (VCO). The VCO receives the signal and sends a specific frequency at a fixed voltage to the variable gain amplifier.
Before operation, the phacoemulsification probe is calibrated by applying a constant voltage to the probe and sweeping the drive circuit through a series of frequencies. Then, a different voltage is selected and another frequency sweep is
performed. This process is repeated for one or more voltage levels and the information on the power and phase versus frequency is stored in memory so that the optimal phase angle at resonance associated with a certain power requirement may be determined
easily, although the phase angle may be relatively constant over a range of power levels. In addition, when the power and phase information is stored in memory, a range of frequencies about a certain resonant frequency is used to create a window beyond
which certain frequencies may not be used.
In operation, a foot pedal is depressed providing a power command which is compared against the existing power. The difference between these two levels is transmitted to the power loop controller. Acting upon the information stored in memory,
the power loop controller selects the appropriate voltage level necessary to correct the difference between the power and the power command and sends this information to the control input of the variable gain amplifier. The variable gain amplifier sends
its output to a power amplifier. The output of the power amplifier is applied to the transformer and simultaneously to both the power monitor and the phase detector. The power is then calculated and compared against the power command signal received
from the foot control and the power loop begins again. The phase detector sends its phase information to a summer/difference block which compares the actual phase against a calculated phase command. The difference between the phase command and the
existing phase is then sent to the frequency loop controller which communicates a signal to the voltage controlled oscillator to emit a certain frequency to the input of the variable gain amplifier which completes the frequency loop. The phase command
is determined from the information taken at calibration time and from the current power command.
Further features and advantages of the present invention, as well as the structure and operation of various embodiments of the present invention, are described below in detail with reference to the accompanying drawings.
BRIEF DESCRIPTION
OF THE DRAWINGS
The accompanying drawings, which are incorporated in and form a part of the specification, illustrate the embodiments of the present invention and together with the description, serve to explain the principles of the invention. In the drawings:
FIG. 1 illustrates a block diagram of an electrical model of an ultrasonic phacoemulsification probe operating near its resonance frequency;
FIG. 2 is a graph of apparent power in accordance with the electrical model of FIG. 1;
FIG. 3 is a graph of the phase angle between the voltage and current waveforms relating to the apparent power graph of FIG. 2 and resulting from the electrical model of FIG. 1;
FIG. 4 is a graph of real power in accordance with the electrical model of FIG. 1;
FIG. 5 is a graph of the phase angle between the voltage and current waveforms relating to the real power graph of FIG. 4 and resulting from the electrical model of FIG. 1;
FIG. 6 is a graph of apparent power and phase angle with the addition of a compensating inductor to the electrical model of FIG. 5;
FIG. 7 is a graph of real power and phase angle with the addition of a compensating inductor to the electrical model of FIG. 5;
FIG. 8 illustrates a block diagram of the phacoemulsification probe system of the present invention;
FIG. 9 illustrates a more detailed apparent power block diagram of the power monitor block in FIG. 8;
FIG. 10 illustrates a more detailed real power block diagram of the power monitor block in FIG. 8;
FIGS. 11, 12, 13, 14 and 15 illustrate a hardware-implemented embodiment of the present invention depicting a coprocessor and an electronically programmable logic device;
FIGS. 16, 17, 18 and 19 illustrate a hardware-implemented embodiment of the present invention depicting memory for the coprocessor and a reset circuit;
FIGS. 20, 21 and 22 illustrate a hardware-implemented embodiment of the present invention depicting a transceiver, and a neuron integrated circuit chip;
FIGS. 23, 2425 and 26 illustrate a hardware-implemented embodiment of the present invention depicting a boost regulator, a voltage controlled oscillator, a multiplying digital to analog converter, a variable gain amplifier, a power amplifier, a
first coupling capacitor, an isolating transformer, a second coupling capacitor, a compensating inductor, and an ultrasonic transducer;
FIGS. 27 and 28 illustrate a hardware-implemented embodiment of the present invention depicting voltage and current RMS to DC converters, and an average power detector;
FIGS. 29 and 30 illustrate a hardware-implemented embodiment of the present invention depicting various minor hardware aspects; and
FIGS. 31 and 32 illustrate a hardware-implemented embodiment of the present invention depicting various minor hardware aspects.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Referring to the accompanying drawings in which like reference numbers indicate like elements, FIG. 8 shows the phacoemulsification probe system, shown generally at 1411, of the present invention. Phacoemulsification probe system 1411 comprises
power loop shown generally at 1412, frequency loop shown generally at 1413, and isolated transducer circuit shown generally at 1414.
As shown in FIG. 8, power loop 1412 comprises power loop controller 1415, variable gain amplifier 1416, power amplifier 1417, first coupling capacitor 1418, transformer 1436, power monitor 1419, first summer/difference block 1425, and power
command signal input 1426.
Power loop controller 1415 has an output to variable gain amplifier 1416. The function of power loop controller 1415 is twofold: (1) a perform a square root operation (power is proportional to the square of the voltage); and (2) to ensure loop
stability and ensure desired system response characteristics. Optionally, the power loop controller 1415 can store in memory peak power information, although this can also be handled by a coprocessor and coprocessor memory combination. Power amplifier
1417 receives an input from the output of variable gain amplifier 1416. The output from power amplifier 1417 proceeds through coupling capacitor 1418 which compensates for leakage inductance as well as blocks any direct current from power amplifier
1417. Power is then delivered to primary transformer 1436 and thence to isolated transducer circuit 1414. In addition, the voltage and current applied to the isolated transducer circuit 1414 are sensed by power monitor 1419. Power monitor 1419
generates a signal proportional to the power (real or apparent).
As shown in FIG. 9, power monitor 1419 may be an apparent power monitor which comprises a voltage root mean square (RMS) to DC converter 1420, current RMS to DC converter 1421, and multiplier 1422. A DC signal providing an apparent power value
is produced which is then communicated to first summer/difference block 1425. Alternatively, power monitor 1419 may be a real power monitor which comprises a voltage and current multiplier 1423 connected to low pass filter 1424. A real power value is
produced with is then communicated to first summer/difference block 1425.
First summer/difference block 1425 compares the power level detected by power monitor 1419 and the power command provided at power command signal input 1426. In hardware, any summer/difference block discussed herein may be embodied as a
difference amplifier, and in software is commonly referred to as a "subtraction" operation. The results of the comparison are communicated to power loop controller 1415. A calculation is made on the magnitude of correction required, and power loop
controller 1415 sends a new signal to voltage gain amplifier 1416 based on the calculation. The calculation may be performed by power loop controller 1415, or any other component associated with power loop controller 1415 such as a coprocessor and
coprocessor memory. This completes one round of power loop 1412.
Frequency loop 1413 comprises frequency loop controller 1430 which communicates a signal to voltage controlled oscillator 1431 which itself provides an input to variable gain amplifier 1416, thence to power amplifier 1417, through coupling
capacitor 1418, to isolated transducer circuit 1414. The phase of the voltage and current waveforms applied to the isolated transducer circuit 1414 are sensed by phase detector 1432 and then communicated to second summer/difference block 1433. A phase
command which is determined from the initial calibration of the system and possibly from subsequent calculation is also communicated to phase command input 1434 of second summer/difference block 1433. Thereafter, second summer/difference block 1433
communicates an error signal based on the phase difference between the actual phase and the phase command to frequency loop controller 1430. A calculation is made on the magnitude of correction required, and frequency loop controller 1430 sends a new
signal to voltage controlled oscillator 1431 based on the calculation. The calculation may be performed by frequency loop controller 1430, or any other component associated with power loop controller 1430 such as a coprocessor and coprocessor memory.
This completes one iteration of frequency loop 1413.
Turning now to isolated transducer circuit 1414, isolated transducer circuit 1414 comprises isolating secondary transformer 1436, second coupling capacitor 1437, compensating inductor 1438, and ultrasonic transducer 1439. More specifically, the
parallel combination of ultrasonic transducer 1439 and compensating inductor 1438 is connected in series with the secondary of transformer 1436 and coupling capacitor 1437. The function of second coupling capacitor 1437 is to compensate for any leakage
inductance from isolating secondary transformer 1436.
The value of compensating inductor 1438 is selected so that the magnitude of its reactance equals the magnitude of the reactance (C) of the parallel capacitance of ultrasonic transducer 1439. If F represents the resonant frequency of | | |
