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BACKGROUND OF THE INVENTION
The present invention relates to a motor control device and power control
device.
Generally, power conversion devices are driven by Pulse Width Modulation
(hereinafter referred to as "PWM") signals obtained by comparing a voltage
command value and a fixed frequency triangular-shaped carrier wave and are
configured to generate a prescribed voltage.
Further, motors can be made small when they are driven by a power
conversion device having a high frequency output current. However, in this
case, beat phenomena oscillating at a low frequency can occur in the
output current of power conversion device, because the frequency of output
current is in the region of 1/10 to 1/3 of the carrier frequency of the
power conversion device.
In a known method of resolving this problem, the frequency of the carrier
is set at a multiple of the output frequency of the power conversion
device, (particularly, an odd number multiple, such as 1, 3, 5, 9), and
the motor is driven with the PWM signal in synchronism with the output
frequency. (This is referred to in the following as the "synchronous PWM
method").
This is disclosed in, for example, Japanese Patent Laid-open Publication
No's. Hei. 7(1995)-227085, Hei. 7(1995)-67350 and Hei. 6(1994)-197550.
These are a first group of example publications where V/F fixed control is
carried out so that a ratio of motor voltage and frequency is
approximately constant.
Moreover, in "HIGH PERFORMANCE VECTOR CONTROLLED THREE-LEVEL GTO INVERTER
SYSTEM FOR ELECTRIC TRACTION" listed in Proceedings of 1995 International
Power Electronics Conference (IPEC-Yokohama '95) a method is shown where
torque of an "alternating current" (hereinafter abbreviated to "a.c.")
motor is controlled using current control while carrying out a synchronous
PWM method using space vectors. This method is capable of current control
up to higher frequencies when compared with the first group of example
publications. This is referred to as a second publication group.
With the synchronous PWM method where carrier waves are compared, when the
carrier wave frequency is an even multiple of the output frequency, a
distorted wave shape of even harmonic waves becomes superimposed with the
output voltage. A third publication example, Japanese Patent Laid-open
Publication No. Hei. 6(1994)-197547, discloses well known technology for
resolving this problem in synchronous PWM methods employing carrier wave
comparisons. Here, the occurrence of even harmonic voltages is suppressed
while making the carrier frequency an even-number multiple of the output
frequency, by performing inversion or non-inversion control on the PWM
signal based on the voltage phase. A finer pulse number can therefore be
achieved in the synchronous PWM method, and improvements in the output
current waveform can therefore be made.
However, these example publications did not take into account the following
points.
The first example publication group, utilizes open loop control of current
and speed based on V/F fixed control, and is therefore not suitable for
applications in products requiring fine control of motor torque and speed
at high speeds. Also, while the carrier frequency for generating the
synchronous PWM signal is clearly decided using a speed command value,
this becomes complex when carrying. out feedback control and can therefore
not be applied as is in this case.
The second example publication, uses a PWM signal generating method
employing space vectors, and complicated operations for PWM signal
generation therefore have to be carried out in an extremely short time. It
is therefore necessary to employ a high-performance microprocessor or a
digital signal processor capable of high speed arithmetic processing at
the control device, which makes the control device expensive.
In the third example publication, a PWM signal-generating processor has to
be added to take into consideration dead time for arm short-circuit
prevention in order to obtain a PWM signal for driving switching elements
of an upper arm and a lower arm of the electric conversion device, which
makes the circuit more complex.
SUMMARY OF THE INVENTION
A first object of the present invention is to provide a motor-control
device capable of controlling output currents of a power conversion device
at frequencies in excess of 1/10 of the frequency of a carrier wave, in a
highly effective manner and with a low-cost. control device.
A second object of the present invention is. to suppress the occurrence of
even harmonic voltages where the carrier wave frequency is an even
multiple of an output frequency at a power control device, while at the
same time obtaining a PWM signal with a simple arithmetic device taking
into consideration dead time for arm short-circuit prevention.
In order to achieve the first object, according to the present invention
there is provided a motor control method for comparing a current of an
a.c. motor and a current command value, calculating an a.c. voltage
command value applied to the a.c. motor from results of the comparison,
generating a pulse width modulation signal by comparing the a. c. voltage
command value and a carrier wave and applying an a.c. voltage to the a.c.
motor using th e pulse width modulation signal in such a manner as to
control the a.c. motor. Here, operation timing of the a.c. voltage command
value is in synchronism with the carrier wave, and the frequency of the
carrier wave is N times (where N is an integer) the frequency of the a.c.
voltage command value.
In order to achieve the second object, according to the present invention
there is provided a motor control system comprising a carrier wave
generator, pulse width modulation controller, an inverter and a switcher.
The carrier wave generator is for generating first and second carrier
waves having a prescribed difference. The pulse width modulation
controller is for generating first and second pulse width modulation
signals by comparing a voltage command value and the first and second
carrier waves. The inverter is for driving a pair of positive side and
negative side switching elements with the first and second PWM signals and
generating a voltage. The switcher is for alternately applying the first
and second pulse width modulation signals to the positive side and
negative side switching elements.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a view showing a configuration of a synchronous motor drive
system of a first embodiment of the present invention;
FIG. 2 is a circuit diagram showing the relationship of a main circuit of
an inverter 3 and a PWM signal;
FIG. 3 is a vector diagram showing the relationship of positions of
magnetic poles of a synchronous motor and voltage vectors;
FIG. 4 is a flowchart showing processing of an interrupt task 1 activated
by an interrupt pulse P1;
FIG. 5 is a flowchart showing processing of an interrupt task 2 activated
by an interrupt pulse P2;
FIG. 6 is a flowchart of a mode setting operation carried out at a mode
setting part 15 in the embodiment of FIG. 1;
FIG. 7 is a characteristic drawing showing the relationship of motor speed
w, decided by the mode setting operation of FIG. 6 and a switching mode;
FIG. 8 is a flowchart of an operation, voltage setting 1, carried out by a
voltage setting part 14 when an interrupt pulse P1 occurs in the
embodiment of FIG. 1;
FIG. 9 is a flowchart of an operation, voltage setting 2, carried out by a
voltage setting part 14 when an interrupt pulse P2 occurs in the
embodiment of FIG. 1;
FIG. 10 is a timing diagram of a synchronous PWM signal when switching mode
Sw=9;
FIG. 11 is a timing diagram showing the relationship of the transfer period
and voltage command value when the phase and period of the output voltage
change;
FIG. 12 is a liming diagram of the synchronous PWM signal when the
switching mode Sw=3;
FIG. 13 is a view showing the configuration of the mode for driving the
synchronous motor using switching mode Sw=6 in a further embodiment of the
present invention;
FIG. 14 is a flowchart showing a mode setting operation carried out at the
mode setting part 15 in the embodiment of FIG. 13;
FIG. 15 is a characteristic drawing showing the relationship of a motor
speed .omega..sub.m decided by the operation of FIG. 14 and the switching
mode;
FIG. 16 is a flowchart of an operation, voltage setting 1, carried out at
the voltage setting part 14 when an interrupt pulse P1 occurs in the
embodiment of FIG. 13;
FIG. 17 is a flowchart of an operation, voltage setting 2, carried out at
the voltage setting part 14 when an interrupt pulse P2 occurs in the
embodiment of FIG. 13;
FIG. 18 is a circuit diagram showing a configuration of a PWM signal
switching circuit 16; and
FIG. 19 is a timing diagram for a synchronous PWM signal when the switching
mode Sw=6 in the embodiment of FIG. 13.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
First Embodiment
The following is a description using FIG. 1 of a first embodiment of the
present invention.
FIG. 1 shows a motor driving device that converts energy of a power supply
2 at a three-phase inverter 3 and drives a synchronous motor 1.
The inverter 3, which is controlled by PWM signals Pu, Pv and Pw outputted
from a control device 4, generates a three-phase a.c. voltage and drives
the synchronous motor 1.
At the control device 4, in order to obtain the PWM signals Pu, Pv and Pw,
motor speed .omega..sub.m of the synchronous motor 1 (from a speed
detector 5), magnetic pole position .theta..sub.m (from a magnetic pole
position detector 6) and phase currents iu and iv (from a.c. sensors 7a
and 7b) are taken as input, and a torque command value tr (outputted from
a torque command generating device 8) is taken in. This control device 4
mainly comprises a vector control operation device 9, a voltage operation
device 10, a PWM generation device 11 and a carrier wave generating device
12.
At the vector control operation device 9 taking the torque command value tr
as input, the torque generated by the motor is obtained by computing a
d-axis current command value idr and a q-axis current command value iqr in
such a manner as to achieve the torque command value tr, taking the motor
speed .omega..sub.m as a parameter. This technique is well known as an
"a.c. motor vector control method", with the d-axis expressing an axis
coincident with the rotational magnetic flux of the a.c. motor and the
q-axis being an axis electrically orthogonal to this axis.
The voltage operation device 10 carries out current control in such a
manner that a d-axis current id and a q-axis current iq become equal to
the d-axis current command value idr and the q-axis current command value
iqr, respectively, and outputs voltage command values Vur, Vvr and Vwr of
phase U, phase V and phase W to the PWM generation device 11. The period
of a carrier wave is also decided at the voltage operation device 10 and a
periodic signal T is outputted to the carrier wave generating device 12.
The carrier wave generating device 12 outputs carrier waves Fcp and Fcn for
the next one period based on the periodic signal T. The carrier waves Fcp
and Fcn are used for generating PWM signals which switch an upper arm and
a lower arm of the inverter 3. Further, interrupt signals P1 and P2 are
generated at the time of the maximum value and minimum value of the
carrier wave Fcp (Fcn) and a function exists for applying the interrupts
in order to activate operations with respect to the voltage operation
device 10.
The PWM generation device 11 calculates PWM signals Pup, Pvp and Pwp of
each phase for the upper arm by comparing the voltage command values Vur,
Vvr and Vwr and the carrier wave Fcp, and calculates PWM signals Pun, Pvn
and Pwn of each phase for the lower arm by comparing the voltage command
values Vur, Vvr and Vwr and the carrier wave Fcn.
The inverter 3 is driven by these PWM signals. The relationship between
switching elements Sup, Svp, Swp, Sun, Svn and Swn of this inverter 3 is
shown in FIG. 2. This operation sequence is described in detail later. The
synchronous motor 1 is driven at a prescribed torque using this
configuration.
Next, the details of the voltage operation device 10 are described. The
voltage operation device 10 is comprised of a current controller 13, a
voltage setting part 14 and a mode setting part 15.
First, the current controller 13 obtains current in a direction coinciding
with the magnetic pole position, i.e. d-axis current id, and current in a
director orthogonal to this d-axis current id, i.e. g-axis current iq from
the phase currents iu and iv using the magnetic pole position qm. Next,
the difference between the d-axis current command value idr and the d-axis
current id, and the difference between the q-axis current command value
iqr and the q-axis current iq are calculated, a current control operation
is carried out, and a d-axis voltage command value Vdr and a q-axis
voltage command value Vqr are obtained. A voltage vector command absolute
value Vr and a voltage vector command phase .theta..sub.m are then
obtained from these command values. The relationship between these values
is shown in the voltage vector diagram of FIG. 3.
The voltage vector command absolute value Vr and the voltage vector command
phase .theta..sub.m are obtained from the vector diagram of FIG. 3 from
the magnetic pole position .theta..sub.m, d-axis voltage command values
Vdr and the qoaxis command value Vqr.
The voltage operation device 10 initiates operationst using the interrupt
signals P1 and P2 as described above, but the operations using the
interrupt signal P1 are processed in the order of the current control
operation of the current controller 13, the mode setting operation of the
mode setting part 15 and the operation voltage setting 1 of the voltage
setting part 14. A current control operation in synchronism with the
carrier wave can therefore be performed by carrying out current control
using this kind of interrupt.
In such a case, motor torque can be generated while carrying out current
control even when the output voltage is in high frequency regions by
introducing an algorithm that puts the carrier wave in synchronism with
the output voltage.
Details of the mode setting operation and the operation voltage setting 1
are shown in the flowcharts of FIG. 6 and FIG. 8. The operation when an
interrupt signal P2 is generated is taken to be just the operation voltage
setting 2 of the voltage setting part 14 as shown in FIG. 5. A flowchart
for the operation voltage setting 2 is shown in FIG. 9. As a result, the
current control operation is carried out once and the voltage setting
operation is carried out twice for one period of the carrier wave.
Sampling periods for the current control and the voltage setting therefore
become one time and 1/2 times the carrier wave period, respectively.
The following is a description of the respective flowcharts.
The mode setting operation shown in FIG. 6 is for deciding the number of
times switching operations are carried out per period of the output
voltage using the motor speed .omega..sub.m. If a switching mode Sw is 0,
an asynchronous PWM method taking the carrier wave frequency to be fixed
is selected regardless of the frequency of the output voltage. The carrier
wave frequency is referred to as a "reference frequency" in this
asynchronous PWM method. When Sw=n (where n#0), a frequency of n-times the
frequency of the carrier wave is set with respect to the frequency of the
output voltage.
A description of the method for deciding this switching mode is described
in FIG. 6. Step 101 branches to step 102 to 106 depending on the motor
speed .omega..sub.m being from low speed to high speed.
When the motor speed .omega..sub.m is less than a speed .omega..sub.1,
where sufficiently stable current control can be carried out even with an
asynchronous PWM method, S2 is taken to be 0 in step 102.
When the motor speed .omega..sub.m is an extremely high speed of speed
.omega..sub.4 or more where the carrier wave frequency cannot be made
high, Sw is decided to be 3 in step 106.
If it is not a PWM method, preferred current control cannot be carried out,
but if the motor speed is within an intermediate range such that the
carrier frequency can be made approximately 9 times the frequency of the
output frequency, i.e. .omega..sub.2 <.omega..sub.m <.omega..sub.3, Sw is
set to 9 in step 104.
In the case of the range of the intermediate motor speed, i.e.
.omega..sub.1 <.omega..sub.m <.omega..sub.2 and .omega..sub.3
<.omega..sub.m <.omega..sub.4, the operations of step 103 and 105 are
carried out so as to give hysteresis due to the switching mode Sw of the
previous time.
Expressing this as a graph, the relationship between the motor speed
.omega..sub.m and the switching mode Sw is as shown in FIG. 7. In doing
so, frequent changing over of the switching mode Sw can be prevented and
smooth transitions can be achieved.
Next, the operation voltage setting 1 carried out at the voltage setting
part 14 is described using the flowchart of FIG. 8.
The operation for the carrier wave period T is first carried out based on
the switching mode Sw obtained at the mode setting part 15. In order to do
this, the switching mode Sw is determined in step 110, with the process
branching to steps 11, 112 or 113 for Sw=0, 9 or 3, respectively.
When Sw=0, the carrier wave period T is put to a fixed value T0 in step
111. As the motor speed .omega..sub.m is comparatively low at this time it
is not necessary to consider beat phenomena in the current control system
and an asynchronous PWM method is acceptable. As the sampling period the
voltage setting is 1/2 of the carrier wave period T, the sampling phase
.theta..sub.s compensating for the sampling time can be obtained from
.omega..sub.m (T/2).
In this respect, when Sw=9, one period of the output voltage is divided by
9 and it is necessary to obtain a carrier wave for generating the PWM
signal.
In step 112, whether or not a current voltage vector command phase
.theta..sub.v is of a prescribed phase that is the output voltage divided
by 9 is obtained using a compensation phase .theta..sub.c. If the voltage
vector command phase .theta..sub.v is n.pi./9 (where n is an integer), a
PWM signal taking a maximum value of a sinusoidal output as center is
symmetrical but the PWM signals for the upper arm and lower arm can be
taken to be the same waveforms with a phase difference therebetween of 180
degrees. Whether .theta..sub.v is (2m+1).pi./9 (where m is an integer).
If the compensation phase .theta..sub.c is 0 in this operation, the
existence of a PWM signal of the original period is indicated. When this
is not the case, the compensation phase .theta..sub.c can be operated upon
in such a manner as to gradually approach 0.
When the.compensation phase .theta..sub.c is greater than .pi./9, 2 .pi./9
is subtracted from .theta..sub.c so as to give a negative value. In doing
the above, the sampling phase .theta..sub.s is reduced by .theta..sub.c /K
(where K is an integer set to a value of, for example, 4 to 50) from an
original .pi./9 in such a manner that the compensation phase .theta..sub.c
gradually approaches zero.
Generation of a carrier wave in synchronism with the phase of a voltage
command value in order to carry out current feedback control is generally
difficult but current control can be achieved using a synchronous PWM
method employing a carrier wave by gradually correcting phase shifts.
Compensation depending on the timing of the taking in of the magnetic pole
position .theta..sub.m is not carried out here but time-divided phase can
be compensated as necessary. As a sampling phase .theta..sub.s is decided
beforehand is step 112, the carrier wave period T can be obtained using 2
.theta..sub.s /.omega..sub.m based on this sampling phase .theta..sub.s.
Similarly, when Sw=3, the voltage vector command phase .theta..sub.v can be
taken to be .pi./3 (where n is an integer) in step 113. Operations are
then carried out in such a manner as to give 2m.pi./3 (where m is an
integer) and the sampling phase .theta..sub.s and the carrier wave period
T are calculated.
Next, in step 114, an output phase .theta. is obtained as the sum of half
of the voltage vector command phase .theta..sub.v, a sampling. phase
.theta..sub.f for the previous time, and the sampling phase .theta..sub.s
for the current time.
As .theta..sub.v is a value based on the magnetic pole position
.theta..sub.m for when the calculation starts and .theta..sub.f is the
phase shift until the results of this calculation are actually set, it is
necessary to add these values. However, as it is necessary for the average
voltage for the next sampling time to coincide with the results of the
current calculation, this object can be achieved by further adding
.theta..sub.s /2.
It is also necessary to correct the voltage command value using the carrier
wave period T. Because the method using a counter is most effective with a
digital circuit, when the carrier wave generating device 12 makes the
carrier wave period T large, the amplitude of the carrier wave also
becomes large. It is therefore necessary to change the voltage vector
command absolute value Vr relative to the carrier wave period T using the
equation V=Vr (T/T0).
An offset voltage Voff required for comparisons with the carrier wave is
also calculated from V0 (T/T0) (where V0 is a reference offset voltage
when T=T0). The voltage command values Vur, Vvr and Vwr for each phase are
therefore obtained.from the equations shown in step 114 of FIG. 8.
On the other hand, a flowchart of the operation voltage setting 2 of the
voltage setting part 14 carried out using the interrupt signal P2 is shown
in FIG. 9.
There is no current control operation carried out here and the carrier wave
period T therefore has the same value as when an operation is carried out
using the interrupt signal P1, and the sampling phase .theta..sub.s can
simply be added to the output phase .theta..
The voltage command values Vur, Vvr and Vwr for each phase are calculated
based on this value. .theta..sub.s is therefore taken to be .theta..sub.f
so that the sampling phase .theta..sub.s for the current time becomes
equal to the sampling phase .theta..sub.f for the previous time when the
next interrupt signal P1 occurs.
In the above, a description is given from the point of view of system
configuration, with synchronous PWM method current control being achieved
while using a carrier wave in this operation sequence.
This operation will now be described in more detail based on time changes
using a timing diagram of FIG. 10.
FIG. 10 is a timing diagram for the switching mode Sw=9. The U-phase
voltage command value Vur is a maximum when the voltage vector command
phase .theta..sub.v is 0 (phase .theta..sub.4 (time t4) but at this time,
the phases of the voltage command Value and the, carrier wave are in
synchronism so that the carrier wave Fc (meaning Fcp and Fcn occurring in
FIG. 1) becomes a minimum. This is then achieved by the compensation phase
.theta..sub.c converging to zero as described by the flowchart of FIG. 8.
The interrupt signals P1 and P2 are respectively generated when the carrier
wave Fc is a maximum and minimum value. Interrupt task 1 and interrupt
task 2 are then activated by these interrupt signals P1 and P2 and the
process described in FIG. 4 and FIG. 5 is carried out.
For example, the carrier wave period T obtained in task 1 processed using
the interrupt signal P1 generated at phase .theta..sub.1 (time t1) is
calculated and set to the carrier wave generating device 12 at the time of
a phase .theta..sub.2 (time t2). At the same time, the voltage command
values Vur, Vvr and Vwr are also set to the PWM generation device 11 and
reflected to the PWM signal for a section from phase .theta..sub.2 (time
t2) phase .theta..sub.3 (time t3).
Because of this, as described in FIG. 8, the sampling phase .theta..sub.f
of the previous time for compensating the phase from activation of the
input task until the voltage command value is set, and 1/2 of the sampling
phase Qs of the previous time for compensating the phase from the voltage
command value being set until actual coincidence with the average voltage
of the converter are added to the voltage vector command phase
.theta..sub.v so as to give the output phase .theta..
The operation depending on the interrupt signal P2 generated at the phase
.theta..sub.2 (time t2) is as follows. The interrupt task 2 is started and
the process of FIG. 5, i.e. the process of FIG. 9, is carried out. The
voltage command values Vur, Vvr and Vwr are then set to the PWM generation
device 11 at the phase .theta..sub.3 (time t3) and reflected at the PWM
signal for the section from phase .theta..sub.3 (time 3) to the phase
.theta..sub.4 (time t4)
The operation results obtained in the interrupt task 1 of phase
.theta..sub.1 (time t1) are used as is at the carrier wave period T as
there are no calculations performed using the interrupt task 2.
The PWM signals Pup, Pvp and Pwp for each of the phases shown in FIG. 10
can be obtained using this process.
By adopting this system configuration, the load on the software processing
required for arithmetic processing for current control etc. can be reduced
while the voltage waveforms for each phase can be made close to being
sinusoidal waveforms using Pwm signals. Control devices can therefore be
made using inexpensive microprocessors with few ripples occurring in any
resulting control system.
FIG. 11 is a timing diagram showing the details of the relationship of the
voltage vector command phase .theta..sub.v from the results of the current
control calculations and the carrier waves Fcp and Fcn, voltage command
value Vur and carrier wave period T when the period of the output voltage
is changed.
The carrier waves Fcp and Fcn are triangular waves usually having a fixed
difference xd for guaranteeing the dead time td for the PWM signal. A
maximum voltage setting value Vmax is then set by the carrier wave
generating device 12 taking into consideration the carrier wave period T.
The carrier waves Fcp and Fcn are incremented or decremented every fixed
time period. For example, when incrementing proceeds for the carrier wave
Fcn so that there is coincidence at a time t1 with the maximum voltage
setting value Vmax, the interrupt signal P1 is generated. Decrementing
then continues until Fcn becomes 0 and the operation for interrupt task 1
is made to start.
The interrupt signal P2 is then generated at a time t2 where the carrier
wave Fcn becomes 0 and a carrier wave period T calculated in the interrupt
task 1 is inputted to the carrier wave generating device 12. The maximum
voltage setting value Vmax is then set based on this carrier wave period
T. The maximum voltage setting value Vmax is then set from the carrier
wave period T using a proportional operation because the actual carrier
wave period is proportional to the maximum voltage setting value Vmax.
At this time, the voltage command values (only Vur of which is shown in
FIG. 11) for each phase set simultaneously at the PWM generation device 11
are taken to be values taking into consideration the maximum voltage
setting value Vmax reflecting changes in the carrier wave period T and the
offset voltage Voff.
As described above, the configuration is such that PWM control is carried
out with extremely fine changes in the carrier wave period T in a
transitionary state of current control and with PWM signals being
generated in synchronism with the phase of the output voltage in a steady
state. Synchronous PWM control can therefore be achieved using carrier
waves while carrying out high-frequency current control.
FIG. 12 is a timing diagram for when the switching mode Sw=3. As in the
case where Sw=9 shown in FIG. 10, the phase of carrier waves Fcp and Fcn
are put into synchronism with the phases of the voltage command values
Vur, Vvr and Vwr by carrying out the process of FIG. 8.
When controlling using the operation of step 113 of FIG. 8, the carrier
wave values are a maximum when the voltage vector command phase
.theta..sub.v is 0, 2.pi./2 and 4.pi./3. The average voltage of the PWM
signal can therefore be made large in a smooth manner in accordance with
increases in the magnitude of the voltage command values.
In this embodiment, an output current can be controlled to be a prescribed
sinusoidal waveform even at frequencies higher than 1/10 that of the
carrier wave frequency using an inexpensive control device that uses
carried waves to generate PWM signals so that a motor control device of
superior torque response can therefore be provided.
Further, if the carrier frequency is the same, the frequency can be made
higher than the highest output frequency of the inverter so that with
permanent magnetic-type synchronous motors in particular, the size of the
magnet used can be reduced, and the costs can be reduced accordingly.
Second Embodiment
FIG. 13 is a view of an embodiment of a motor control device for
controlling current using a synchronous PWM signal where the carrier wave
frequency is an even multiple (6 times) of the output frequency of the
inverter. Here, the point of distinction with the first embodiment of FIG.
1 is that a PWM signal switching circuit 16 is added in FIG. 13. The
contents of the processing carried out at the voltage setting part 14 and
the mode setting part 15 within the voltage operation device 10 is also
modified with respect to FIG. 1.
First, the process of the flowchart shown in FIG. 14 is carried out in
place of the process of FIG. 6 at the mode setting part 15. In FIG. 14, a
process for providing at a section where Sw=6, i.e. a step 107 and step
108, is added between the switching modes Sw=3 and Sw=9. In this way, as
shown in FIG. 15, Sw=6 within the range of .omega..sub.m from
.omega..sub.4 to .omega..sub.5. Further, the switching mode Sw is decided
for within the ranges .omega..sub.3 to .omega..sub.4, and .omega..sub.5 to
.omega..sub.6 while maintaining a hysteresis characteristic.
In the first embodiment, the high-frequency component of the waveform for
the output current rapidly changes by three times while th switching mode
Sw switches from 9 to 3 because the switching frequency is reduced by 1/3.
However, the amount of change of the high-frequency component is reduced
in this second embodiment by introducing a switching mode Sw=6 at the
center.
FIG. 13 differs from FIG. 1 in that the switching mode Sw=6 is inputted at
the voltage setting part 14 and in that voltage signal signals Fu, Fv and
Fw for each phase are outputted.
The process of the voltage setting part 14 is carried out during the
interrupt tasks 1 and 2 activated by the interrupt signals P1 and P2, with
the contents of the respective processes being shown in FIG. 16 and FIG.
17.
FIG. 16 differs from FIG. 8 in that in that voltage signal signals Fu, Fv
and Fw for each phase are set to 1 in step 114 and in that step 115 and
116 are added.
Here, the values 1 and 0 for the voltage signal signals Fu, Fv and Fw mean
"switching" and "no switching", respectively. In the step 114, the
switching mode Sw is 3 or 9. The voltage signal signals Fu, Fv and Fw are
set to 1 as with the first embodiment of FIG. 8.
A description will now be given for when Sw is determined to be equal to 6
in step 110 and the processing of step 115 and 116 is carried out.
The compensation phase .theta..sub.c is obtained in steps 115 and 116 in
the same way as in step 112 and step 113 in order to generate a carrier
wave where one period of the output voltage is divided by 6, with the
sampling phase .theta..sub.s then being corrected using the compensation
phase .theta..sub.s so that the compensation phase .theta..sub.c
approaches 0. The process for calculating the carrier wave period T,
output phase .theta., corrected voltage V and offset voltage Voff from the
sampling phase .theta..sub.s is the same as for steps 113 and 114.
The adding of the offset voltage Voff to absolute values for sine waves for
the voltage command values Vur, Vvr and Vwr is also a point of difference
with the other switching modes.
Further, the voltage signal signals Fu, Fv and Fw are set to one when the
sine wave sign is positive and 0 when the sine wave sign is negative.
The same processing is also carried out in the operation (voltage setting
2) of the voltage setting part processed in interrupt task 2 shown in FIG.
17 as is described for the case of switching mode Sw=6 in FIG. 16.
The voltage signal signals Fu, Fv and Fw obtained in this way are compared
at the PWM generation device 11 as shown in FIG. 13, outputted as
fundamental PWM signals Pu1, Pu2, Pv1, Pv2, Pw1 and Pw2 for each phase and
inputted to the PWM signal switching circuit 16. The PWM signal switching
circuit 16 is comprised of a U-phase processor 17u, V-phase processor 17v
and a W-phase processor 17w, as shown in FIG. 18, and carries out the same
processing.
A description is given here of the U-phase processor 17u. Delay circuits
18a and 18b that delay, rising by a prescribed dead, time only on the
rising part of the signal are provided at an input part for the voltage
signal Fu. With the circuits provided, there is no delay for when a signal
falls.
With the above circuit configuration, Pu1 is outputted as the PWM signal
Pup and Pu2 is outputted as the PWM signal Pun when the voltage signal Fu
is 1, i.e., when the voltage sign is positive. On the other hand, Pu1 is
outputted as PWM signal Pun and Pu2 is outputted as PWM signal Pup when Fu
is 0, i.e. when the voltage sign is negative.
This is expressed as waveforms in a timing diagram in FIG. 19. Here, Pu1 is
a usual wide band PWM signal and Pus is a usual narrow band PWM signal.
The manner in which a U-phase upper arm PWM signal Pup and a lower arm PWM
signal Pun are obtained from the relationship of output signals Fu1 and
Fu2 coming from the voltage signal Fu via the delay circuits 18a and 18b
and a NOT circuit and signals Pu1 and Pu2 can be understood from the
circuit diagram of FIG. 18 and the timing diagram of FIG. 19.
The signals Pup and Pun are the same waveforms but are 180 degrees out of
phase with each other. The occurrence of even harmonic frequencies in the
output current of the inverter 3 can therefore be suppressed as a result
of using these waveforms. If the PWM generating method for switching mode
Sw=6 is taken to be the same as for Sw=3 and 9, even harmonic frequencies
occur because Pup and Pun are totally different waveforms.
In this second embodiment, upper arm and lower arm PWM signals can be
simultaneously obtained while taking dead time into consideration and
circuit configuration can therefore be simplified. It is further also
possible to take a microcomputer having a dedicated circuit for generating
a PWM signal using two carrier waves currently on the market (for example,
SH7034 of Hitachi Co. Ltd. ) and further simplify circuit configuration
with this embodiment.
It is also possible to manufacture an LSI dedicated to inverter control
employing a synchronous PWM signal by incorporating the PWM signal
generating circuit included in circuits up to FIG. 18 into a
microcomputer.
In the embodiments described here, the waveforms for the PWM signals Pup
and Pun are not strictly symmetrical taking the phase 0 as center because
delay circuits 18a and 18b that only delay on rises of signals are used
but symmetrical waveforms can be obtained by modifying the configuration
of the circuit.
According to these embodiments, smooth synchronous PWM control switching
can be carried out even for motor control | | |
