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BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a control device which controls a
synchronous motor including a reluctance motor, and a method of
controlling a synchronous motor including a reluctance motor, and an
electric motor vehicle control device using the same.
2. Conventional Art
In order to control such as speed and torque of a synchronous motor it is
necessary to detect or estimate its magnetic pole position, and thus the
speed and torque of the synchronous motor can be controlled through a
current control or a voltage control thereof based on the detected or
estimated magnetic pole position.
Conventionally, the magnetic pole position was detected by a position
detector. However, recently a method of controlling a synchronous motor
while estimating the magnetic pole position, in that a control method with
magnetic pole position sensorless has been proposed which is different
from the conventional method of detecting the magnetic pole position by
making use of a position sensor.
For example, Takeshita, Ichikawa et al. "Control of Salient Type Brushless
DC Motor with Sensorless Based on Estimation of Speed Electromotive Force"
(Collected Papers of Japanese Electrical Engineers Society Vol. 117-D,
No.1, 1997) proposes a method of performing speed control of a motor while
estimating a speed electromotive force by making use of a motor model.
Further, JP-A-8-205578 (1996) discloses a method of detecting a salient
pole characteristic of a synchronous motor based on a correlation of
ripple components of a voltage vector applied to the synchronous motor
through a pulse width modulation control (hereinafter referred to as PWM
control) and of the corresponding motor current vector.
The art disclosed in the above paper is a method of estimating the magnetic
pole position based on a difference between a current calculated on the
control model and an actual motor current flowing therethrough, and has a
feature that a control system can be formed only through control
calculations in a controller.
Further, since the art disclosed in JP-A-8-205578 (1996) uses general PWM
signals which control a voltage of the synchronous motor, the method has
an advantage that no additional signals for detecting the magnetic pole
position are required.
Further, with the method of estimating magnetic pole position based on a
difference between a current calculated from a control model and an actual
motor current flowing therethrough, there was an unsolved problem that
once the synchronous motor steps out on any causes, the synchronous motor
may be brought into an out-of-control condition.
On the other hand, with the art disclosed in JPA-8-205578 (1996) the
magnetic pole position of the synchronous motor can always be detected by
its salient pole characteristic, therefore, the synchronous motor is never
brought into an out-of-control condition.
However, with the method of detecting the magnetic pole position of a
synchronous motor through its salient pole characteristic, it is necessary
to detect a correlation between the motor current state and the applied
voltage every time when the PWM signals change.
Namely, it is necessary to detect the motor current state and to grasp the
applied voltage state at least six times for one cycle of a carrier wave,
for this reason there arose a problem that the calculation speed can not
catch up with, if a controller of high performance is not used.
SUMMARY OF THE INVENTION
An object of the present invention is to provide a synchronous motor
control device which can be produced with low cost.
Another object of the present invention is to provide a highly reliable
synchronous motor control system.
One of the measures according to the present invention is to calculate,
namely to estimate a magnetic pole position of the synchronous motor based
on a variation amount or a variation direction of a motor current when the
synchronous motor is put in a short circuited state and to control the
synchronous motor based on the calculated magnetic pole position.
Other measures according to the present invention will be explained in the
Detailed Description of the Preferred Embodiments.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram showing one embodiment of the present invention
in which a magnetic pole position of a cylinder type synchronous motor is
detected by making use of a current differential circuit;
FIG. 2 is a circuit diagram of the inverter 3 in FIG. 1;
FIG. 3 is a time chart showing a relation between a carrier wave signal,
three phase voltage command values and PWM signals, and a fetching timing
of an inverter current in the embodiment shown in FIG. 1;
FIG. 4 is a flow chart when detecting magnetic pole position in the
embodiment shown in FIG. 1;
FIG. 5 is a block diagram showing another embodiment of the present
invention in which the magnetic pole position is calculated by detecting
motor currents when two phases of a cylinder type synchronous motor is
under a short circuited condition;
FIG. 6 is a time chart showing a relation between a carrier wave signal,
three phase voltage command values and PWM signals, and a fetching timing
of an inverter current in the embodiment shown in FIG. 5;
FIG. 7 is a flow chart when detecting magnetic pole position in the
embodiment shown in FIG. 5;
FIG. 8 is a Table showing arithmetic expressions for calculating current
difference values when two phases being short circuited and phases of
current differential vectors when three phases being short circuited in
steps 115 and 116 in FIG. 7;
FIG. 9 is a block diagram showing still another embodiment of the present
invention in which the magnetic pole position of a salient type
synchronous motor is detected by making use of the inverter currents while
prolonging the three phase short circuited interval;
FIG. 10 is a time chart showing a relation between a carrier wave signal,
three phase voltage command values and PWM signals, and a fetching timing
of an inverter current in the embodiment shown in FIG. 9;
FIG. 11 is a flow chart when detecting magnetic pole position with a high
accuracy in the embodiment shown in FIG. 9;
FIG. 12 is a block diagram showing a further embodiment of the present
invention which comprises a magnetic pole position sensor for controlling
a salient type synchronous motor for an electric motor vehicle and a
magnetic pole position detecting means which detects the magnetic pole
position thereof based on the inverter currents when two phases being
short circuited;
FIG. 13 is a flow chart when detecting the magnetic pole position of the
salient pole type synchronous motor by making use of the inverter currents
when two phases being short circuited in the embodiment shown in FIG. 12;
FIG. 14 is a Table showing arithmetic expressions for calculating current
difference values when two phases being short circuited and phases of
current differential vectors when three phases being short circuited in
steps 136 and 137 in FIG. 13;
FIG. 15 is a flow chart when performing an abnormality judgement of the
magnetic pole position in the embodiment shown in FIG. 12;
FIG. 16 is a block diagram showing a still further embodiment of the
present invention which includes a self diagnosis function of false
detection in magnetic pole position in a salient pole type synchronous
motor having a magnetic pole position detecting means detecting the
magnetic pole position by making use of inverter currents when two phases
being short circuited;
FIG. 17 is a vector diagram showing an exemplary relation between a current
vector, a differential current vector and magnetic pole position, in other
words, d axis in a synchronous motor;
FIG. 18 is a vector diagram showing a relation between a differential
current vector when two phases being short circuited and a differential
current when three phases being short circuited in the cylinder type
synchronous motor shown in FIG. 9;
FIG. 19 is a vector diagram showing a relation between differential current
vectors which are generated by a voltage applied on a axis of a salient
pole type synchronous motor;
FIG. 20 is a vector diagram showing a relation between a differential
current vector when two phases being short circuited and a differential
current when three phases being short circuited in the salient pole type
synchronous motor shown in FIG. 16;
FIG. 21 is a diagram of a synchronous motor control system showing another
embodiment according to the present invention;
FIG. 22 is a diagram showing a possible region in which detection accuracy
of the magnetic pole position reduces;
FIG. 23 is a diagram showing a structure of a calculating unit 52 in FIG.
21;
FIG. 24 is a diagram showing a structure of a current command value
generating unit 6 in FIG. 21;
FIG. 25 is a diagram showing a magnetic characteristic of a synchronous
motor;
FIG. 26 is a diagram showing a d axis characteristic of a synchronous
motor;
FIG. 27 is a flow chart showing a processing sequence for detecting a
magnetic pole position during the time when a synchronous motor is
started; and
FIG. 28 is a flow chart showing a processing sequence for discriminating
polarity of magnetic pole based on torque generating direction and
rotating direction of a rotor shaft of a synchronous motor.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Hereinbelow, an embodiment of the present invention will be explained with
reference to FIG. 1.
FIG. 1 is a block diagram of a motor control device in which a cylinder
type synchronous motor 1 is driven by DC energy from a battery 2. The DC
voltage of the battery 2 is inverted by an inverter 3 into a three phase
AC voltage, which is applied to the cylinder type synchronous motor 1. The
inverter 3 is controlled based on an output of a controller 4.
The output of the controller 4 is determined based on the following
calculation result. Although the controller 4 in FIG. 1 is illustrated in
a functional block diagram, the controller 4 can be realized not only by a
hardware but also by a software. A differential circuit 12, a current
detector unit 10 as well as a PWM signal generating unit 9, which will be
explained later, use partly an input/output circuit of a computer. For
example, the input/output circuit is such as an analogue/digital converter
and a pulse output circuit, and through their use all of the functions can
be performed by software programs.
Namely, at first a current command value generating unit 6 determines a d
axis current command value idr and a q axis current command value iqr with
respect to a torque command value .tau.r to be generated from the motor 1.
Further, the torque command value .tau.r is issued to the current command
value generating unit 6 from a control device or a control program which
is in a higher hierarchy with respect to the controller 4.
The d axis is a direction in the magnetic pole or the magnetic fluxes, the
q axis is electrically orthogonal direction to the d axis, and d axis and
q axis in combination constitute d-q axes coordinate system. When a rotor
with magnets of a motor rotates, the d-q axes coordinate system also
rotates, therefore, a phase of the d-q axes coordinate system from a
stationary coordinate system, in that .alpha.-.beta. axes coordinate
system, is assumed as .theta.. Namely, an object of the present embodiment
is to detect the phase .theta. of the magnetic pole (hereinbelow, referred
to as magnetic pole position .theta.) based on inverter currents.
FIG. 17 shows a vector diagram illustrating one exemplary relation between
coordinate systems and currents therein. If the d axis current and the q
axis current can be controlled according to the command values, the
synchronous motor 1 can generate a torque coincident with the torque
command value .tau.r. The value of the torque command .tau.r is commanded
either directly to the current command value generating unit 6 or
indirectly via a speed control calculating circuit (not showing). Signals
respecting the values of a U phase current iu and a V phase current iv
from current sensors 5a and 5b are sent to a current detecting unit 10 and
are detected by the current detecting unit 10 at a current detection
timing P1 which will be explained later. The detected current values are
respectively converted by a coordinate system converting unit 11 into a d
axis current id and a q axis current iq for the d-q axes coordinate
system.
In the present embodiment, the currents detected by the current detecting
unit 10 are two phase currents iu and iv of U phase and V phase, this is
because W phase current iw can be determined by the U and V phase currents
iu and iv and the detection of W phase current iw is omitted. Of course,
all of the three phase currents can be detected.
A current control unit 7 calculates a d axis current deviation between the
d axis current command value idr and the d axis current id and a q axis
current deviation between the q axis current command value iqr and the q
axis current, and performs a proportion and integration calculation for
the respective deviations to determine a d axis voltage command value Vdr
and a q axis voltage command value Vqr.
A voltage setting unit 8, which receives the d axis voltage command value
Vdr and the q axis voltage command value Vqr, calculates three phase
voltage command values Vur, Vvr and Vwr for the stationary coordinate
system based on a magnetic pole position .theta. and outputs the same to a
PWM signal generating unit 9.
The PWM signal generating unit 9 calculates three phase PWM pulses Pup,
Pvp, Pwp, Pun, Pvn and Pwn and outputs the same to the inverter 3.
FIG. 2 shows a relation between the circuit connection diagram of the
inverter 3 and the PWM pulses therefor. For example, when the PWM pulse
Pup is high, a switching element Sup is turned on, and when the Pup is
low, the switching element Sup is turned off.
Further, the PWM pulses Pup and Pun are generally in an opposite relation
with regard to high and low state. However, in order to prevent a power
source short circuiting, a short circuit preventing interval is provided
which keeps the both PWM pulses in a low state, when the state of the PWM
pulses are inverted.
Processing contents performed in the PWM signal generating unit 9 are
explained with reference to a timing chart as shown in FIG. 3. Through
comparison of the wave forms of the respective phase voltage command
values Vur, Vvr and Vwr with triangular wave shaped carrier waves, three
phase PWM pulses Pup, Pvp and Pwp are obtained. Further, an illustration
of the above mentioned short circuit preventing interval is omitted in the
drawing for simplify the explanation.
Namely, when the PWM pulses Pup, Pvp and Pwp are in high in FIG. 3, the
switching elements Sup, Svp and Swp in upper arms in FIG. 2 are
respectively placed in an on state, and the switching elements Sun, Svn
and Swn in lower arms therein are respectively placed in an off state.
When the PWM pulses Pup, Pvp and Pwp are low, the switching elements Sun,
Svn and Swn are respectively in an on state and the switching elements
Sup, Svp and Swp are respectively in an off state.
As will be seen from FIG. 3, when the voltage command values of the
respective phases are in a predetermined range including maximum value and
minimum value of the carrier waves, there exists an interval in which
three phases either in the upper arms or in the lower arms are in a short
circuited condition. When the detection use pulse P1 is designed to be
generated when the carrier wave reaches to its maximum value and to its
minimum value, the detection use pulse P1 is resultantly generated when
the three phases of the synchronous motor are in a short circuited state.
Further, it is known that when the current detection unit 10 is designed to
detect the currents of the respective phases when the detection use pulse
P1 is generated, the detected instantaneous current values substantially
correspond to respective average current values of the concerned phases.
Still further, the short circuited state of respective phase windings in
the synchronous motor exists not only at a moment of the maximum value and
the minimum value of the carrier waves as shown in FIG. 3 but also exists
in a predetermined range including the same. The predetermined range is
represented by a pulse width among PWM pulses Pup, Pvp and Pwp having the
narrowest pulse interval and by an interval between the most wide pulse
and the adjacent pulse thereto. Timing t1 appears in a width range of
pulse Pvp, timing t2 appears between two successive pulses Pup, timing t3
appears in a width range of pulse Pvp, timing t4 appears between another
successive two pulses Pup, timing t5 appears in a width range of pulse Pwp
and timing t6 appears between still another two successive pulses Pup.
Still further, the timings t1 through t6 represent moments either the
maximum value or the minimum value of the carries waves. As has been
explained above during a predetermined interval including the moments of
the respective maximum and minimum values the short circuited state of the
phase winding is caused and which is repeated. In order to take out a
current flowing through the windings under a short circuited state
thereof, the pulse p1 is produced. It is sufficient when the pulse p1 is
generated at the predetermined interval. The method according to the
present embodiment in which the detection use pulses are generated at the
timings of the maximum value and the minimum value of the carrier waves
shows advantages such as that the detection use pulses are easy to produce
and a possibility of erroneous operation is reduced, because the detection
use pulses are generated at substantially the center period of the short
circuited state.
Now, an important principle of the present embodiment as shown in FIG. 1 is
explained.
A current differential circuit 12 is inputted of signals representing such
as the U phase current iu and the V phase current iv and outputs
differential current values piu and piv obtained by differentiating or
affine differentiating the input current values.
These differentiated current values such as piu and piv are inputted into a
detection unit 13 and are held until the detection use pulse P1 is
generated, and thereafter are outputted. Namely, the current differential
values piu and piv are detected at the timing of the pulses p1, in other
words are fetched into a calculating unit 14.
The calculating unit 14 which calculates a magnetic pole position performs
the processings as shown in the flow chart in FIG. 4 to determine the
magnetic pole position .theta..
At first, in step 101 the differentiated current values piu and piv when
the three phases are short circuited, are inputted into the calculating
unit 14.
In step 102, a phase .gamma. of a differentiated current vector pis, when
the three phases are short circuited, is calculated and determined.
In FIG. 17, phase relations of the differentiated current vector pis with
respect to other vectors are illustrated. From the differentiated current
values piu and piv when the three phases are short circuited an .alpha.
axis differentiated current value pi.alpha. and a .beta. axis
differentiated current value pi.beta. can be determined.
When the U phase axis coincides with the .alpha. axis, the .alpha. axis
differentiated current value pi.alpha. and the .beta. axis differentiated
current value pi.beta. are respectively obtained by the following
arithmetic formulas;
pi.alpha.=(3/2)piu (1)
pi.beta.=(1/2)(piu-2piv) (2)
Subsequently, the phase .gamma. is calculated based on the thus determined
values pi.alpha. and pi.beta. by making use of the relations illustrated
in FIG. 17.
In step 103, the magnetic pole position .theta. is determined according to
the following arithmetic formula;
.theta.=.gamma.+.pi./2 (3)
One of the feature of the present embodiment is our discovery that a
relation between the magnetic pole position .theta. and the phase .gamma.
of the three phase short circuited current is approximately expressed by
the above arithmetic formula (3) of which ground will be explained below.
Fundamental operation of a synchronous motor in d-q axes coordinate system
are expressed by the following arithmetic formulas, wherein p=d/dt and
.omega. represents a rotating angular speed of the motor;
Vd=(R+pLd)id-.omega.Lqiq (4)
Vq=(R+pLq)iq+.omega.(Ldid+.phi.) (5)
When a synchronous motor is placed in a three phase short circuited state,
the applied voltage of the synchronous motor stands Vd=Vq=0, therefore,
the condition of the synchronous motor when the three phases are short
circuited is expressed by the following arithmetic formulas;
pid=(.omega.Lqiq-Rid)/Ld (6)
piq=-{.omega.(Ldid+.phi.)+Riq}/Lq (7)
The differentiated current vector in the stationary .alpha.-.beta. axes
coordinate system is a sum of the differentiated current vector in d-q
axes coordinate system and a differentiated current vector generated
through the rotation of the d-q axes coordinate system at an angular speed
.omega., therefore, a d axis differentiated current value pids and a q
axis differentiated current value pigs seen from the .alpha.-.beta. axes
coordinate system are respectively expressed by the following arithmetic
formulas;
pids={.omega.(Lq-Ld)iq-Rid}/Ld (8)
piqs=-{.omega.(Ld-Lq)id+.phi.)+Riq}/Lq (9)
Accordingly, the phase .delta. of the differentiated current vector when
three phase are short circuited with respect to d axis, namely the
magnetic pole position .theta., is expressed by the following arithmetic
formula;
tan
(.delta.)=piqs/pids=-Ld[.omega.{(Ld-Lq)id+.phi.}+Rid]/
[Lq{.omega.(Lq-Ld)iq-Rid}] (10)
In the present embodiment, since the cylinder type synchronous motor 1 is
used, a condition Ld=Lq is given, therefore, the above arithmetic equation
(10) is modified as follows;
tan (.delta.)=Ld(.omega..phi.+Riq)/(LqRid) (11)
When id<0, the phase .delta. is approximated by the following arithmetic
formula;
.delta..apprxeq.-.pi./2 (12)
For this reason, the calculation according to the arithmetic formula (3) is
performed in step 103.
When the motor angular speed .omega. is low, the error based on the
approximation (12) increases, therefore, the phase .delta. can be obtained
asymptotically based on the arithmetic formula (11) of which method will
be explained later in connection with other embodiments.
As has been explained above, through a simple calculation in the
calculating unit 14 as shown in FIG. 1 the magnetic pole position .theta.
can be determined. When coordinate conversions are performed in the
voltage setting unit 8 and in the coordinate conversion unit 11 based on
the thus determined magnetic pole position .theta., the motor is
controlled to generate a required torque corresponding to a torque command
value.
Accordingly, the present embodiment is characterized by the fact that the
magnetic pole position of a cylinder type synchronous motor can be
detected through a comparatively simple calculation only with the
provision of current sensors without using a mechanical magnetic pole
position sensor such as a resolver and encoder which directly measures the
rotating position of the magnetic pole of the cylinder type synchronous
motor. For this reason the control device is produced with a low cost.
Further, even if the synchronous motor steps out on any causes, the
synchronous motor is never brought into an out-of-control condition,
because the magnetic pole position can always be detected.
Moreover, the present embodiment is characterized by the fact that in
parallel with a usual PWM control since a sensorless control system is
constructed only by making use of information obtained during the
performance of the PWM control, noises and torque ripple of the
synchronous motor are reduced in comparison with the conventional method
of detecting the magnetic pole position by applying detection use
additional signals.
FIG. 5 is a block diagram of another embodiment for a cylinder type
synchronous motor in which the magnetic pole position is detected without
using a current differential circuit. Like FIG. 1 embodiment, the present
embodiment is also realized not only by electric circuits but also by
computer softwares.
Major different points of the present embodiment from that shown in FIG. 1
embodiment are that the current differential circuit 12 is not used, the
current detection timing is modified by an introduction of a detection use
pulse P2 and a different processing other than that in the calculating
unit 14 as shown in FIG. 1 is performed in the calculating unit 15. An
important feature of the present embodiment is that the three phase short
circuited current is not directly detected.
Now, the detection use pulse P2 which controls detection timing of the
current detector unit 10 is explained with reference to FIG. 6. FIG. 6
shows the same state of PWM signals as that shown in FIG. 3, however, the
current detection use pulses P2 as shown in FIG. 6 is different from the
current detection use pulses P1 as shown in FIG. 3 in the following
points.
With respect to respective phases of a 180.degree. conduction type three
phase inverter as shown in FIG. 2, either the switching element in the
upper arm or the switching element in the lower arm is usually placed in
an on state and the other is placed in an off state. For this reason, at
least two phases among the three phases are always short circuited.
FIG. 6 illustrates such interval. For example, in a time section from time
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